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Electrical Engineering
Biljana Milivojevic
Study of Optical Differential Phase Shift Keying
Transmission Techniques at 40 Gbit/s and
beyond
PH.D. Dissertation
Paderborn, April 2005
DISSERTATION
ON
Study of Optical Differential Phase Shift Keying
Transmission Techniques at 40 Gbit/s and
beyond
SUBMITTED IN PARTIAL FULFILLMENT OF THE
REQUIREMENTS FOR THE DEGREE OF
DOKTOR-INGENIEUR
IN ELECTRICAL ENGINEERING
(Dr.-Ing.)
TO
DEPARTMENT OF ELECTRICAL ENGINEERING
UNIVERSITY OF PADERBORN
WARBURGER STR. 100, 33098 PADERBORN
GERMANY
BY
Dipl.-Ing. Biljana Milivojevic
from Serbia and Montenegro
Reviewers:
1. Prof. Dr.-Ing. Reinhold No´
e
2. Prof. Dr.-Ing. Andreas Thiede
Date of Thesis Submission: April 7, 2005
Date of Defense Examination: June 22, 2005
Paderborn, April 2005
Diss.
Dedicated to
my parents, my husband and my dearest kid Nikola
Contents
Contents V
List of Figures VIII
List of Tables XIII
List of Publications XV
ABSTRACT XIX
1 Introduction 1
1.1 Background.................................. 1
1.2 Motivation................................... 3
1.2.1 oDPSK and oDQPSK Transmission . . . . . . . . . . . . . . . . . 3
1.2.2 High-Speed Integrated Circuits for xPSK Transmission . . . . . . . 4
1.3 OrganizationofThesis ............................ 5
2 oDPSK Transmission System 7
2.1 oDPSK signal generation . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
2.1.1 Differential Encoding and Decoding . . . . . . . . . . . . . . . . . 8
2.1.2 Optical Phase Modulation . . . . . . . . . . . . . . . . . . . . . . 11
2.1.3 40 Gbit/s (CS)RZ-DPSK transmitter . . . . . . . . . . . . . . . . . 16
2.1.4 Experimental Results . . . . . . . . . . . . . . . . . . . . . . . . . 21
2.2 oDPSKsignaldetection............................ 21
2.2.1 Mach-Zehnder Interferometer Modelling . . . . . . . . . . . . . . 22
2.2.2 40 Gbit/s RZ-DPSK receiver . . . . . . . . . . . . . . . . . . . . . 23
2.2.3 Measurement results . . . . . . . . . . . . . . . . . . . . . . . . . 25
2.3 Signed On Line Chromatic Dispersion Detection . . . . . . . . . . . . . . 27
2.3.1 Chromatic Dispersion in Single Mode Fibers . . . . . . . . . . . . 27
2.3.2 Measurement Setup for Chromatic Dispersion Detection . . . . . . 28
2.3.3 Experimental results . . . . . . . . . . . . . . . . . . . . . . . . . 29
2.4 Chromatic Dispersion Compensation . . . . . . . . . . . . . . . . . . . . . 31
2.4.1 Adaptive Tunable CD Compensation . . . . . . . . . . . . . . . . 33
V
2.5 Conclusion .................................. 38
3 oDQPSK Transmission System 39
3.1 Introduction to oDQPSK . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
3.2 oDQPSK signal generation . . . . . . . . . . . . . . . . . . . . . . . . . . 41
3.2.1 DQPSKPrecoding.......................... 41
3.2.2 OpticalEncoder ........................... 42
3.2.3 40 Gbaud DQPSK Transmitter . . . . . . . . . . . . . . . . . . . . 47
3.3 oDQPSK signal detection . . . . . . . . . . . . . . . . . . . . . . . . . . . 48
3.3.1 DQPSKDecoding .......................... 48
3.3.2 40 Gbaud DQPSK Receiver . . . . . . . . . . . . . . . . . . . . . 49
3.4 2×40 Gbit/s DQPSK Transmission Experiment . . . . . . . . . . . . . . . 49
3.4.1 Transmission setup . . . . . . . . . . . . . . . . . . . . . . . . . . 49
3.4.2 Measurement Results and Discussion . . . . . . . . . . . . . . . . 50
3.5 RZ-DQPSK Polarization Multiplex Transmission . . . . . . . . . . . . . . 53
3.5.1 Transmission Setup . . . . . . . . . . . . . . . . . . . . . . . . . . 53
3.5.2 Transmission Results . . . . . . . . . . . . . . . . . . . . . . . . . 55
3.6 Conclusion .................................. 58
4 High-Speed Integrated Circuits for oDPSK Transmission 59
4.1 Differential Amplifier for 10 and 40 Gbit/s CS(RZ)-DPSK system . . . . . 59
4.2 Differential Distributed Amplifier . . . . . . . . . . . . . . . . . . . . . . 60
4.2.1 Distributed Amplification . . . . . . . . . . . . . . . . . . . . . . 61
4.2.2 CircuitDesign ............................ 63
4.2.3 ResultDisscusion........................... 68
4.3 10 Gbit/s CMOS Differential Amplifier . . . . . . . . . . . . . . . . . . . 68
4.3.1 Design of Transmission Line Structures . . . . . . . . . . . . . . . 69
4.3.2 CircuitDesign ............................ 74
4.3.3 ResultDisscusion........................... 77
4.4 Conclusion .................................. 77
5 Result Discussion and Future Scope 79
5.1 DPSKTransmission.............................. 79
5.2 DQPSKTransmission............................. 80
5.3 High-Speed Integrated Circuit for oDPSK Transmission . . . . . . . . . . . 81
5.4 Conclusion .................................. 82
A Definitions 83
A.1 BitErrorRate................................. 83
A.2 Receiversensitivity.............................. 84
A.3 Optical signal-to-noise ratio . . . . . . . . . . . . . . . . . . . . . . . . . 84
B Theory of the Traveling Wave Amplifier 87
VI
C Extraction of Transmission Line Parameters 93
References 96
Acknowledgements 107
VII
VIII
List of Figures
1.1 TrafficGrowthTrends............................. 2
2.1 Schematic diagram of a differentially encoded (left) and decoded (right) bit 9
2.2 Proposal for regular differential encoding/decoding scheme with realizable
feedbackdelays................................ 9
2.3 Simplified differential encoding/decoding scheme . . . . . . . . . . . . . . 10
2.4 Realizedscheme ............................... 10
2.5 Principle of optical PSK signal modulation a) data signal b) carrier c) PSK
signal ..................................... 11
2.6 Lithium Niobate-based Phase Modulator . . . . . . . . . . . . . . . . . . . 12
2.7 Waveguidebasedtravelling-wavephasemodulatorusingx- or z-cut LiNbO3
materials.................................... 13
2.8 X-cut Lithium Niobate-based Mach-Zehnder modulator . . . . . . . . . . . 14
2.9 Three different structures for Mach-Zehnder modulator using x- or z-cut
LiNbO3.................................... 15
2.10 Chromaticdispersiontolerance of DPSK using either a Mach-Zehnder mod-
ulator or a phase modulator at the data rate of 40 Gbit/s and chromatic
dispersion of 0, 34, and 68 ps/nm . . . . . . . . . . . . . . . . . . . . . . . 16
2.11 40 Gbit/s CSRZ-DPSK transmitter . . . . . . . . . . . . . . . . . . . . . . 17
2.12 Photograph of the data buffer board . . . . . . . . . . . . . . . . . . . . . 17
2.13 DPSK Transmitter using the MZMs . . . . . . . . . . . . . . . . . . . . . 19
2.14 DPSK signal generation . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
2.15 a)Transmission characteristic of MZM b)Optical intensity generated signals 20
2.16DSPKopticalreceiver ............................ 21
2.17 SimplifiedInterferometermodel with delay T and retardation R inthelower
branch..................................... 22
2.18 Block diagram of the lock-in amplifier’s scheme . . . . . . . . . . . . . . . 25
2.19 Optical spectrum at the constructive port (left) and destructive port(right) . 25
2.20 BER vs. power at optical preamplifier input for different CS-RZ DPSK
modulationformat .............................. 26
2.21 40 Gbit/s eye diagrams back-to-back for NRZ-DPSK (left) and (CS)RZ-
DPSK(right) ................................. 26
IX
2.22 Total dispersion Dand relative contributions of material dispersion Dmand
waveguide dispersion Dwfor a conventional single mode fiber . . . . . . . 28
2.23 Experimental 40 Gbit/s CSRZ-DPSK setup for chromatic dispersion detec-
tion ...................................... 29
2.24 Chromatic dispersion detection readout vs. actual dispersion. Inset: eye
diagram resulting from interferometer output signal difference . . . . . . . 30
2.25 Standard deviation versus measurement interval, at zero actual dispersion . 30
2.26 CSRZ-DPSK eye diagrams at interferometer outputs (top), and difference
signal (bottom) back to back (left) after transmission over the 91km (right) . 31
2.27 Illustration of a uniform grating with constant amplitude of refractive index
modulation and grating period . . . . . . . . . . . . . . . . . . . . . . . . 32
2.28 Principle of FBG CD compensator with circulator . . . . . . . . . . . . . . 33
2.29 CDC Setup for 40 Gbit/s DPSK transmission . . . . . . . . . . . . . . . . 34
2.30 Photograph of the TeraXion thermally tunable dispersion compensator . . . 35
2.31 Group delay versus wavelength in tunable CD compensator for dispersion
settings .................................... 35
2.32 OSNR needed for BER = 10−9versus compensator CD . . . . . . . . . . . 36
2.33 BER versus OSNR. The OSNR is varied by an attenuator. . . . . . . . . . . 37
2.34 40 Gbit/s eye diagrams back-to-back (top) and after 263 km transmission
(bottom), for NRZ-DPSK and CSRZ-DPSK (from left to right) . . . . . . . 37
3.1 DQPSKConstellations ............................ 40
3.2 Schematic representation of Optical DQPSK signalling . . . . . . . . . . . 41
3.3 DQPSK signal generation using two Mach-Zehnder modulators . . . . . . 43
3.4 DQPSK signal generation using Mach-Zehnder and phase modulator . . . . 43
3.5 Single dual-drive MZM for DQPSK signal generation . . . . . . . . . . . . 44
3.6 Procedure to find ϕi1and ϕi2for si=riejθi.................. 46
3.7 DQPSK signal generation using a dual-drive Mach-Zehnder modulator and
interferometer................................. 46
3.8 2×40 Gbit/s DQPSK Transmitter . . . . . . . . . . . . . . . . . . . . . . 47
3.9 40Gbaudintensity eyediagramsofNRZ-DQPSK(left)andCS(RZ)DQPSK
signals(right)................................. 48
3.10DQPSKDecoder ............................... 48
3.11 2×40 Gbit/s RZ-DQPSK transmission setup . . . . . . . . . . . . . . . . 50
3.12 Measured BERs vs. optical preamplifier input power for RZ-DPSK, RZ-
DQPSK,RZ-ASK............................... 51
3.13 2×40 Gbit/s RZ-DQPSK I and Q eye diagrams back-to-back (top) and
after 263 km of fiber (middle). Bottom diagram is back-to-back with wrong
interferometerphase ............................. 52
3.14 4×40 Gbit/s per channel RZ-DQPSK PolDM transmission . . . . . . . . . 53
3.15 Electrical interference spectra measured in the 12 GHz photoreceiver after
thepolarizer.................................. 55
X
3.16 Back-to-back receiver sensitivity for both in-phase and quadrature data
channelsforone polarization. Optical powerisgivenforaggregate160 Gbit/s
signal ..................................... 55
3.17 Back-to-back performance of 4×40 Gbit/s system . . . . . . . . . . . . . 56
3.18 Eye diagrams in one polarization, (top) back-to-back in I channel, Q chan-
nel and (bottom )after 230 km in I and Q channel . . . . . . . . . . . . . . 56
3.19 Optical spectrum after 229 km of fiber . . . . . . . . . . . . . . . . . . . . 57
3.20 Measured Q factors for I and Q data channels in both polarizations back-
to-back for 8WDM channels, and after transmission over 230 km fiber for
the CD-compensated 192.5THzchannel................... 57
4.1 Typical 40 Gbit/s CS(RZ)-DPSK balanced optical front end . . . . . . . . . 60
4.2 Simulated DC characteristics of the HEMT fabricated in OMMIC D01PH
process .................................... 61
4.3 Basic configuration of the travelling wave amplifier . . . . . . . . . . . . . 62
4.4 Typical schematic of the cascode amplifier . . . . . . . . . . . . . . . . . . 64
4.5 Schematic of the differential pre-amplifier (left) and simulated magnitude
of S21 andCMRR(right)........................... 64
4.6 Schematic of a traveling wave amplifier using cascode as the main ampli-
fyingstage................................... 65
4.7 Stability factors kand µ(left) and output reflection coefficient S22 (right) . 65
4.8 Phases on the gate and drain line . . . . . . . . . . . . . . . . . . . . . . . 66
4.9 Layout details of the cascode cell . . . . . . . . . . . . . . . . . . . . . . . 66
4.10 Optimization of forward transmission as a function of number of stages N
(left) and group delay (right) . . . . . . . . . . . . . . . . . . . . . . . . . 67
4.11 Simulated eye diagram for 50 mVpp input voltage . . . . . . . . . . . . . . 67
4.12 Layout of the differential distributed amplifier at 40 Gbit/s . . . . . . . . . 67
4.13 Cross-section of the 0.18 µm CMOS process (left) and n-MOS transcon-
ductance as a function of the gate-source voltage (right) . . . . . . . . . . . 69
4.14 Geometry of the microstripline in CMOS (left) and its characteristic im-
pedance as function of the conductor width (right) . . . . . . . . . . . . . . 70
4.15 Geometry of coplanar waveguide . . . . . . . . . . . . . . . . . . . . . . . 70
4.16 Characteristic impedance of the coplanar waveguide in function of the ratio
W/(W+ 2G)(left) and conductor width (right) . . . . . . . . . . . . . . . 71
4.17 Measured attenuation for 50 ΩCPW versus width of signal line . . . . . . 71
4.18 SL configuration (left) and characteristic impedance of the SL in function
of the conductor width (right) . . . . . . . . . . . . . . . . . . . . . . . . . 72
4.19 Microphotograph of the fabricated MS, CPW and SL (from left to right) . . 72
4.20 Comparison of measured MS, CPW and SL data for the magnitude S21 and
S11 ....................................... 73
4.21 Comparison of measured, simulated and modelled MS data for the magni-
tude S21 (top-left), phase S21 (top-right) and magnitude S11 (bottom) . . . . 73
XI
4.22 Comparison of measured, simulated and modelled CPW data for the mag-
nitude S21 (top-left), phase S21 (top-right) and magnitude S11 (bottom) . . . 74
4.23 Comparison of measured, simulated and modelled SL data for the magni-
tude S21 (top-left), phase S21 (top-right) and magnitude S11 (bottom) . . . . 74
4.24 Schematic of the differential amplifier using striplines in CMOS . . . . . . 75
4.25 Microphotograph of the realized chip . . . . . . . . . . . . . . . . . . . . . 75
4.26 Comparison of measured, simulated single phase and simulated differential
magnitude of S21 ............................... 76
4.27 Comparison of measured, simulated single phase and simulated differential
magnitude of S11 ............................... 76
4.28 Measured eye diagram at 10 Gbit/s for 27−1PRBS input signal . . . . . . 77
B.1 Lumped transmission line with shunt loss . . . . . . . . . . . . . . . . . . 87
B.2 One section of the drain line . . . . . . . . . . . . . . . . . . . . . . . . . 89
B.3 One section of the gate line . . . . . . . . . . . . . . . . . . . . . . . . . . 89
B.4 Signal path from the input to the output via the ’k’-th transistor . . . . . . . 91
C.1 Single transmission line represented by a two-port network and described
with distributed transmission line parameters R, L, C and G→0....... 93
XII
List of Tables
1.1 The frequencies of the optical carriers and the propagation losses in single
mode optical fiber in the three most popular optical bands . . . . . . . . . . 1
1.2 Opticalcarrierrates.............................. 2
2.1 Selected long-haul 40 Gb/s DPSK transmission experiments . . . . . . . . 7
2.2 The bit stream to be transmitted and bit stream generated for DPSK trans-
mission .................................... 8
3.1 Phase states for DQPSK signal . . . . . . . . . . . . . . . . . . . . . . . . 40
3.2 Selected DQPSK transmission experiments with higher spectral efficiencies 41
4.1 Distributed circuit parameters for interconnect test structures . . . . . . . . 73
XIII
List of Publications
(1) B. Milivojevic, A. Fauzi Abas, A. Hidayat, S. Bhandare, D. Sandel, R. No´
e, M. Guy,
M. Lapointe,“1.6 bit/s/Hz, 160 Gbit/s, 230 km RZ-DQPSK Polarization Multiplex Trans-
mission with Tunable Dispersion Compensation”, IEEE Photonics Technology Letters, vol.
17 (2005), 495–497.
(2) B. Milivojevic, Z. Gu, A. Thiede, “10 Gbit/s Differential Amplifier Demonstrating
Striplines in 0.18m CMOS Technology”, Proc. European Microwave Week, The European
Gallium Arsenide and other Compound Semiconductors Application Symposium, Paris,
France (October 2005), accepted.
(3) B. Milivojevic, S. Hoffmann, A. Thiede, R. No´
e, R. Leblanc, B. Wroblewski, “Distrib-
uted Amplifiers for Transmitter and Receiver of a 40 Gbit/s DPSK Optical Transmission
System”, Proc. European Microwave Week, The European Gallium Arsenide and other
Compound Semiconductors Application Symposium, Amsterdam, The Netherlands, (Octo-
ber, 2004), 9–12.
(4) B. Milivojevic, A. Fauzi Abas, A. Hidayat, S. Bhandare, D. Sandel, R. No´
e, M. Guy,
M. Lapointe, “160 Gbit/s, 1.6 bit/s/Hz RZ-DQPSK Polarization-Multiplexed Transmission
over 230 km Fiber with TDC”, Proc. European Conference on Optical Communication
ECOC, Stockholm, Sweden, (2004), We1.5.5.
(5) B. Milivojevic, D. Sandel, S. Bhandare, R. No´
e, F. W¨
ust, “Chromatic Dispersion De-
tection in a 40 Gbit/s CSRZ-DPSK Transmission”, Proc. YUINFO, Kopaonik, Serbia and
Montenegro, (2004), CD-58, 91.
(6) B. Milivojevic, D. Sandel, S. Bhandare, R. No´
e, F. W¨
ust,“40 Gbit/s CSRZ-DPSK
Transmission System with Signed Online Chromatic Dispersion Detection”, IEE Electron-
ics Letters, vol. 39 (2003), 1455–1456.
(7) B. Milivojevic, D. Sandel, S. Bhandare, R. No´
e, F. W¨
ust, “Practical 40 Gbit/s CSRZ-
DPSK Transmission System with Signed Online Chromatic Dispersion Detection”, Proc.
European Conference on Optical Communication ECOC, Rimini, Italy, (2003), Tu3.6.4.
(8) A. Fauzi, B. Milivojevic, A. Hidayat, S. Bhandare, D. Sandel, H. Zhang, R. No´
e,
“2.38 Tbit/s, 1.49 bit/s/Hz, 40 Gbit/s RZ-DQPSK Polarization Division Multiplex Trans-
mission over273kmofFiber”, ElectricalEngineeringarchiveforElectrotechnics, Springer-
Verlag GmbH, (July, 2005), ISSN: 0948-7921, 1432-0487 (Online), DOI: 10.1007/s00202-
004-0274-y.
(9) S. Bhandare, D. Sandel, B. Milivojevic, A. Hidayat, A. Fauzi, H. Zhang, S. K. Ibrahim,
F. W¨
ust, R. No´
e, “5.94 Tbit/s (40×2×2×40 Gbit/s) C-Band Transmission over 324 km
XV
using RZ-DQPSK Combined with Polarization Division Multiplex”, 6. ITG-Fachtagung
Photonische Netze, Leipzig, Germany, (May, 2005), 87–90.
(10) S. Bhandare, D. Sandel, B. Milivojevic, A. Hidayat, A. Fauzi, H. Zhang, S. K.
Ibrahim, F.W¨
ust, and R.No´
e, “5.94 Tbit/s, 1.49bit/s/Hz(40×2×2×40Gbit/s)RZ-DQPSK
Polarization Division Multiplex C-Band Transmission over 324 km”, IEEE Photonics Tech-
nology Letters, vol. 17 (April, 2005), 914–916.
(11) D. Sandel, S. Bhandare, B. Milivojevic, R. No´
e, M. Guy, M. Lapointe, “Automated
tunable chromatic dispersion compensation at 40 Gbit/s”, Proc. 5. ITG-Fachtagung Pho-
tonische Netze, Leipzig, Germany, (May, 2004), 199–201.
(12) S. Bhandare, D. Sandel,B. Milivojevic, A. Fauzi Abas Ismail, A. Hidayat, R. No´
e,
“2×40 Gbit/s RZ-DQPSK transmission”, Proc. 5. ITG-Fachtagung Photonische Netze,
Leipzig, Germany, (May, 2004), 195–197
(13) D. Sandel, S. Bhandare, A. Fauzi, B. Milivojevic, R. No´
e, M. Guy, and M. Lapointe,
“Automatic Tunable Chromatic Dispersion Compensation at 40 Gbit/s in ASK and DPSK
NRZ and CSRZ, 263 km Transmission Experiment”, IEEE Photonics Technology Letters,
vol. 16 (2004), 2568–2569.
(14) S.Bhandare, D. Sandel, A. F. Abas, B. Milivojevic, A. Hidayat, R. No´
e, M.Guy,
M. Lapointe, “2×40 Gbit/s RZ-DQPSK transmission with tunable chromatic dispersion
compensation in a 263 km fiber link”, IEE Electronics Letters, vol. 40 (2004), 821–822.
(15) R. No´
e, D. Sandel, S. Bhandare, F. W¨
ust, B. Milivojevic, V. Mirvoda, “ Signed on-
line chromatic dispersion monitoring by synchronous detection of FM-induced arrival time
modulations in the clock recovery PLL”, Journal of Optical Networking, Optical Society
of America, vol. 3 (2004), 589–600.
(16) A. Fauzi, D.Sandel, A. Hidayat, B. Milivojevic, S. Bhandare, H. Zhang, R. No´
e,
“2.56 Tbit/s, 1.6 bit/s/Hz, 40 Gbaud RZ-DQPSK polarization division multiplex transmis-
sion over 273 km of fiber”, Proc. OECC/COIN2004, Yokohama, Japan, (2004), PD1-4.
(17) S.Bhandare, D. Sandel, A. Fauzi, B.Milivojevic, R. No´
e, M. Guy, M. Lapointe, “Fully
automatic, tunable chromatic dispersion compensation at 40 Gbit/s in ASK and DPSK,
NRZ and CSRZ, 263 km transmission experiments”, Proc. OECC/COIN2004, Yokohama,
Japan, (2004), 15C1-4.
(18) D.Sandel, S. Bhandare, A. Fauzi, F. W¨
ust, B. Milivojevic, A. Hidayat, R.No´
e, M. Guy,
M. Lapointe, “2×40 Gbit/s RZ-DQPSK transmission over 263 km of fiber with tunable
XVI
chromatic dispersion compensator”, Proc. OECC/COIN2004, Yokohama, Japan, (2004),
16C2-3.
(19) S. Bhandare, A. Hidayat, D. Sandel, A. F. Abas, H. Zhang, B.Milivojevic, R. No´
e,
M. Guy, M. Lapointe, “Adaptive 700...1350 ps/nm Chromatic Dispersion Compensation in
1.6 Tbit/s (40×40 Gbit/s) DPSK and ASK Transmission Experiments over 44...81 km of
SSMF”, 6. ITG-Fachtagung Photonische Netze, Leipzig, Germany, (May, 2005), 91–94.
XVII
ABSTRACT
The return-to-zero Differential Phase Shift Keying (RZ-DPSK) transmission format has
attracted the interest of transmission experimentalist as an enabler for 40 Gbit/s systems
because it provides 3dB more system margin. Combined with Return-to-Zero (RZ) coding,
it is also more resilient to non-linear distortions imposed by the transmission fiber. Here,
a40 Gbit/s Carrier-Suppressed RZ-DPSK transmission system with receiver sensitivity of
−33.5dBm is presented. Various new features such as demodulation of RZ-DPSK signals
using a delay interferometer having delay of 4symbol durations and lock-in stabilization
of the delay interferometer phase are demonstrated. It is particularly shown that a signed
online chromatic dispersion measurement scheme which synchronously detects arrival time
variations in the clock recovery phase locked loop also works for this type of modulation
formats.
Further more, the residual chromatic dispersion of a various fiber link lengths up to
263 km length is automatically compensated for NRZ-DPSK and CSRZ-DPSK modula-
tion formats at 40 Gbit/s, using synchronous arrival time detection scheme and a fiber
Bragg grating–based thermally tunable dispersion compensator in the range of −300 to
−700 ps/nm. The total measured penalty of transmission and CD compensation is −1.2dB
... +1.2dB, for various link lengths and compensator CDs.
A simple alternative to double the existing transmission capacity without optical band-
width increase is to use Differential Quadrature Phase Shift Keying (DQPSK). Combined
with RZ coding its robustness against cross-phase modulation is also large because the in-
tensity is not modulated by the data. In this work, 2×40 Gbit/s (40 Gbaud) RZ-DQPSK
transmission over 263 km of fiber with manually thermally tunable chromatic dispersion
compensator with a back-to-back Q factor >20 dB (extrapolated BER <10−23) is re-
ported. The receiver sensitivity is −27.5dBm. Even after transmission the Q factor is
17.5dB.
DQPSK and polarization division multiplex (PolDM) transmission each double fiber
capacity by their increased spectral efficiency. Both techniques have been combined to
transmit 4×40 Gbit/s per WDM channel. The fiber capacity is 1.6bit/s/Hz, the value which
has been previously reported or surpassed only at 10 Gbaud. A 1.6bit/s/Hz transmission
over 230 km of fiber is demonstrated with Q>15.6dB for one of the 8WDM channels for
which the thermally tunable dispersion compensator was operational.
A differential amplifier combined with travelling wave amplifier at 40 Gbit/s is designed
and simulated in a pseudomorphic AlGaAs/InGaAs HEMT technology. The gain and 3dB
bandwidth are 17 dB and 46 GHz, respectively.
A version of the amplifier that will work at 10 Gbit/s is simulated, designed, and fab-
ricated in low-cost 0.18µm CMOS technology. Amplifier makes use of striplines. The
experiment demonstrates the 10 Gbit/s signal propagation over narrow CMOS striplines.
For a single phase input, amplifier has a gain of 6dB at 10 Gbit/s. The measured bandwidth
is 6.2GHz and common mode rejection ratio (CMRR) is 8dB. This CMOS circuit using
striplines exhibits a performance comparable performances with that of the state-of-the-
art amplifiers designed in conventional technologies. This opens the possibility for using
XIX
striplines with its over all good shielding in complex analog systems. Such differential in
differential out amplifiers can be used in balanced optical front ends to achieve the record
sensitivity values.
XX
Chapter 1
Introduction
1.1 Background
Concept of optical fiber communication was introduced 20 years ago, when it became
possible to reduce the attenuation in glass fibers to a few dB per kilometer. At that time,
mostfiber optic communication system were using multimodegraded-indexfibersandlaser
source radiated around 850 nm. These systems were mainly dispersion limited. Today’s
optical transmission systems are using the single mode optical fibers. They are operating
in the transmission window where the material dispersion (1310 nm) or the attenuation
(1550 nm) is minimum in single mode optical fibers. In single mode optical fiber, the
light is confined to an area of high refractive index, which acts as core, with respect to its
surroundings, called as clad. The Table 1.1 lists the frequencies of optical carriers and the
propagation losses of the single mode optical fibers in the three most popularly used optical
transmission windows.
Table 1.1: The frequencies of the optical carriers and the propagation losses in single mode optical fiber in
the three most popular optical bands
Wavelength (µm) Frequency (THz) Loss (dB/km)
0.85 353 2.0
1.31 229 0.5
1.55 194 0.2
The optical fiber communication industry also experienced the several generations like
the integrated circuit technology. The digital optical transmission systems characterized by
the available data rates rapidly took over the analog communication systems. The data rate
levels commonly used in the digital fiber optical networks are given in Table 1.2. SONET
refers to Synchronous Optical Networks, a North American standard while SDH refers to
Synchronous Digital Hierarchy, a EU standard. The first generation systems were specified
as OC-1 having the basic data of 51.84 Mb/s. OC stands for the optical carrier and each
succeeding generation has a data rate which is integer multiple of OC-1.
1
2Chapter 1 Introduction
Table 1.2: Optical carrier rates
SONET SDH Data Rate (MB/s)
OC-1 STM-0 51.84
OC-3 STM-1 155.52
OC-12 STM-4 622.08
OC-48 STM-16 2488.32
OC-192 STM-64 9953.28
OC-768 STM-256 39813.12
OC-13072 STM-1024 159252.48
The main driving force behind the development of high bit rate transmission systems
and the future capacity upgrades are expected due to the exponential growth of the inter-
net traffic. According to the prediction of [1], the expected capacity increase caused by
voice based services is almost negligible (4%) compared to other possible broadband ser-
vices (Figure 1.1). A doubling in internet traffic each year appears a more likely outcome,
according to the Moore’s law. Transmission systems using high spectral efficiencies are,
therefore needed to accommodate the expected capacities in a bandwidth limited case.
1994 1998 2002 2006
Year
3.0
2.0
1.0
0.0
Tbit/s
Figure 1.1: Traffic Growth Trends
However, duo to the global economic slow down the predictions from early 90s are no more
valid. Recent analysis of US market [2] predict the annual grow of 10.3% for the fiber-optic
related products and subsystems through out the year 2006.
Commercially available digital optical transmission systems which are fully operational
today mainly suffer from the smaller spectral efficiencies (typically <0.4bit/s/Hz) as they
use the so called Intensity Modulation with Direct Detection (IM/DD) technique. Spectral
efficiency is specified in terms of the specified data rate in Gb/s in 100 GHz band limitation.
1.2 Motivation 3
Intensity modulated systems at higher data rates are also mainly limited by the so called
polarization mode dispersion (PMD) [3] once the linear fiber dispersion (chromatic disper-
sion) is compensated. Therefore, the researchers are looking for the alternative modulation
formats that could be implemented in the future generations of the digital optical trans-
mission systems. The newer alternative modulation formats should directly influence the
spectral efficiencies and increase the fiber capacity thereby satisfying the growing data traf-
fic demands.
1.2 Motivation
The motivation behind this work is to explore the area of modulation formats which are
primarily based on phase shift keying (PSK), as an alternative to the conventional inten-
sity modulated formats such as nonreturn-to-zero (NRZ), return-to-zero (RZ) or carrier
suppressed return-to-zero (CSRZ). Multi-level PSK-based modulation format offers the
possibility to transmit more than one bit of information for the per transmitted optical
symbol. This directly increases the spectral efficiency without the optical bandwidth in-
crease. During this work, the digital optical transmission system at 40 Gbit/s is developed
based on differential phase shift keying (DPSK) and this concept is later extended to dif-
ferential quadrature phase shift keying (DQPSK) modulation format which provides 2:1
multiplexing and 1:2 demultiplexing at the optical level (1.2.1). Such systems indeed need
the high-speed differential amplifiers for balanced detection (1.2.2).
1.2.1 oDPSK and oDQPSK Transmission
Even though PSK sensitivity performance is better than DPSK [4], DPSK is more preferred
because demodulation of PSK requires coherent detection where the synchronous optical
carrier is regenerated in the receiver so that the phase information can be correctly extracted
at the receiver side without any ambiguity. This puts the stringent requirements on the
laser linewidths of the transmitter laser and also on the local oscillator laser. These phase
fluctuation can also be mitigated, for example, by using a variant of PSK with differential
encoding at the transmitter side and a differential decoding at the receiver side. Generally,
RZ or CSRZ-DPSK is used instead of NRZ-DPSK. In RZ- or CSRZ-DPSK, the intensity is
not modulated by the data but is rather modulated by pulse carving. Thus, the information
is carried by the optical phase itself. That’s the reason why this modulation format is
particularly more robust to cross phase modulation [5–7] and to PMD. A detailed review
of such systems using DPSK as an alternative modulation format is given in [8]. The
very first optical systems based on PSK were extensively studied with respect to coherent
detection [9,10]. However DPSK is generally demodulated using incoherent detection, for
example, by a delay line demodulator followed by balanced detection and therefore, yields
3 dB more system margin as compared to conventional NRZ [11,12]. Recently, number of
experiments are reported using DPSK as an alternative and promising modulation format
for long-haul and ultra long-haul optical transmission systems [13–20].
4Chapter 1 Introduction
A practical 40 Gb/s CSRZ-DPSK system with signed online chromatic dispersion de-
tection is developed as a part of this work at the University of Paderborn, and is reported
in [21,22]. Adaptive chromatic dispersion compensation is demonstrated in 263 km trans-
mission experiments, for the first time, for NRZ- and CSRZ-DPSK modulation formats,
using a thermally tunable dispersion compensator based on fiber Bragg grating technol-
ogy [23].
A simple alternative to increase the fiber capacity is to use Differential Quadrature
Phase Shift Keying (DQPSK) [24–28], which doubles the existing transmission capacity
by carrying the two-bits of information for the each transmitted optical symbol [25] with-
out optical bandwidth increase. Combined with RZ coding its robustness against cross
phase modulation (XPM) is also large because the intensity is not modulated by the data.
DQPSK signal also tolerates strong optical filtering [29]. The theoretically possible re-
ceiver sensitivity of DQPSK receiver is better than for intensity modulation. 2×40 Gbit/s
(40 Gbaud) RZ-DQPSK transmission over 263 km fiber is reported in [30]. This work has
demonstrated the sufficient resilience against non-linear phase noise and a band limitation
in a 40 Gbit/s WDM DEMUX with Q factors >17.5dB.
The highly spectrally efficient transmission is the key to cost-effective capacity expan-
sion in optical communication systems with a finite limited bandwidth. DQPSK [31,32]
and polarization division multiplex (PolDM) [33], transmission each double fiber capacity
by their increased spectral efficiency [6,27,33]. 160 Gbit/s (4×40 Gbit/s) Polarization
Division Multiplex RZ-DQPSK transmission over 230 km of fiber with Q > 15.6dB is
demonstrated in one of 8 100 GHz-spaced WDM channels for which a thermally tunable
dispersion compensator was operational [34].
1.2.2 High-Speed Integrated Circuits for xPSK Transmission
The future digital optical transmission systems will either use DPSK or DQPSK modu-
lation formats, as has been mentioned in section 1.2.1. D(Q)PSK receivers are generally
based on balanced detection technique and hence need the balanced optical front ends to-
gether with differential amplifiers. A differential amplifier concept using pseudomorphic
AlGaAs/InGaAs HEMT technology is presented in [35]. The circuit basically consists of
a differential preamplifier followed by Travelling Wave Amplifier’s (TWA’s) as the main
amplifying stages. Traditionally, high-speed circuits were realized either in GaAs [36–39],
InP [40–42] or in SiGe technology. However, all the above mentioned technologies have
relatively high cost of integration. Another alternative is to use the standard CMOS tech-
nology to drastically reduce the cost, especially when the 40 Gbit/s circuits are scaled down
to 10 Gbit/s. In this work, a single-stage differential pre-amplifier followed by three pairs of
common source stages using striplines is implemented in standard 0.18 µm CMOS technol-
ogy. Simulated and measured results are summarized in [43]. CMOS differential amplifier
presented in this work exhibits a performance which is comparable to the state-of-the-art
amplifiers designed in other conventional technologies [44–46].
1.3 Organization of Thesis 5
1.3 Organization of Thesis
The thesis is mainly organized into five chapters. First chapter is nothing more than a brief
introduction to the thesis.
Second chapter deals with the development of a 40 Gbit/s CSRZ-DPSK transmission
system with signed online chromatic dispersion detection. Fully automatic tunable chro-
matic dispersion compensation is demonstrated using signed online chromatic dispersion
detection for DPSK signals. Third chapter describes the generation and transmission of
2×40 Gbit/s CSRZ-DQPSK signals with and without polarization multiplexing with the
thermally tunable manual chromatic dispersion compensation.
Fourth chapter is devoted to design and simulation of high-speed integrated circuits in
GaAs and CMOS technologies. Measurement results on the differential amplifier fabri-
cated in 0.18 µm CMOS technology are also summarized in this chapter. Fifth chapter
presents summary of all successfully conducted transmission experiments along with the
future scope of this work.
6Chapter 1 Introduction
Chapter 2
oDPSK Transmission System
Return-to-zero differential phase shift keying (RZ-DPSK) has received considerable atten-
tion over the last few years as it provides 3 dB improvement in receiver sensitivity using
the balanced detection technique compared to standard NRZ modulation format [11]. This
improvement directly translates into a reduced optical signal-to-noise ratio (OSNR) re-
quirement to achieve a given bit-error-rate (BER), thereby increasing the system margin.
This additional margin can be used to extend the transmission distance, reduce optical
power requirement, or relax component specification. RZ-DPSK is also more robust to
penalties caused by cross-phase modulation (XPM) in multichannel optical transmission
systems [18,47]. A number of 10 Gb/s transmission experiments using RZ-DPSK are re-
ported in [8,17,18,47,48]. Table 2.1 summarizes a few selected 40 Gb/s long-haul trans-
mission experiments recently carried out using DPSK as modulation format. During this
work, a practical carrier-suppressed RZ-DPSK transmission is demonstrated with signed
online chromatic dispersion detection and compensation.
Table 2.1: Selected long-haul 40 Gb/s DPSK transmission experiments
Number of Channels Data Rate Distance Fiber Type Efficiency Reference no.
Channels (Gb/s) (km) (km) (b/s/Hz)
64 42.7 4.000 NZDSF 0.4 [14]
40 42.7 10.000 DM 0.4 [15]
40 42.7 8.740 DM 0.57 [19]
100 43 6.240 DM 0.4, 0.64 [20]
64 42.7 8.200 DM 0.8 [7]
149 42.7 6.120 DM 0.8 [49]
160 42.7 3.200 DM 0.8 [16]
NZDSF: Nonzero dispersion shifted fiber
DM: Dispersion managed fibers
7
8Chapter 2 oDPSK Transmission System
2.1 oDPSK signal generation
Phase shift keying formats carry the information in the optical phase itself. Due to the lack
of absolute phase reference in direct-detection receivers, the phase of the preceding bit is
used as a relative phase reference for demodulation. This results in differential phase shift
keying modulation formats which carry the information in optical phase changes between
bits. The next subsection illustrates how to generate differentially encoded sequence in the
lab environment.
2.1.1 Differential Encoding and Decoding
Generally, differentially encoded bit sequence is obtained from the input binary message
sequence by using
gi=gi−1⊕di,(2.1)
where the index iidentifies the consecutive bits at 40 Gbit/s, diis the binary input infor-
mation bit sequence to the encoder, giis the output bit sequence of the encoder and gi−1
is the 1-bit delayed version of the gi. This symbol ⊕denotes modulo-2 addition. The en-
coded symbol gis transmitted in a bipolar fashion as the optical field polarity as shown in
Table 2.2. The differential encoding requires the use of a reference bit before initiating the
encoding process. This reference bit could be either arbitrarily set to logic “1” or to logic
“0”.
Table 2.2: The bit stream to be transmitted and bit stream generated for DPSK transmission
di0 1 1 1 0 0 1 1 0
gi1 0 0 0 0 1 0 0 0 1
phase 0 π π π π 0πππ0
Thus, differentially encoding is schematically represented by a two-input XOR gate
whose one of the two inputs is driven by the encoding bit sequence, while the other input
is driven by its 1-bit delayed version of the encoded bit sequence. Since 40 Gbit/s bit rate
corresponds to 25 ps bit duration, the differential encoding at 40 Gbit/s can be implemented
as in Figure 2.1 (left). The received symbol is differentially decoded in an interferometer
having a 1-bit delay. When written with binary variables the result is
di=gi−1⊕gi,(2.2)
just as desired. Schematically differential decoding is represented as shown in Figure 2.1
(right).
2.1 oDPSK signal generation 9
25 ps
di
gi gidi
25 ps
gi-1gi-1
Figure 2.1: Schematic diagram of a differentially encoded (left) and decoded (right) bit
New Proposal: Regular Differential Encoding/Decoding Scheme
A practical problem that occurs in DPSK signal generation at 40 Gbit/s is the necessity of
recursive differential encoding at 40 Gbit/s, since physically realizable feedback delays ex-
ceed 25 ps. Here, this problem is taken into consideration and a new, fully regular encoding
scheme (Figure 2.2) is presented. Differential encoding,
ei=ei−4⊕di(2.3)
takes place only at 10 Gbit/s in 4 separate, parallel circuits. Then, the signals are then 4:2
multiplexed to 20 Gbit/s and modulo-2-added in 2 parallel circuits according to
fi=ei⊕ei−2.(2.4)
After the additional 2:1 multiplexing to 40 Gbit/s, a very similar, serial non-recursive oper-
ation occurs
gi=fi⊕fi−1.(2.5)
2:1
MUX
50ps
25ps
1-bit differentially
encoded giat 40Gbit/s
4data
streams
diwith
10Gbit/s
each
2:1
MUX
100ps
100ps
2:1
MUX
100ps
100ps
50ps
1-bit differentially
encoded fiat
20Gbit/s
1-bit differentially
encoded eiat
10Gbit/s
Figure 2.2: Proposal for regular differential encoding/decoding scheme with realizable feedback delays
10 Chapter 2 oDPSK Transmission System
2:1
MUX
4-bit differentially
encoded at 40Gbit/s
4data
streams
diwith
10Gbit/s
each
2:1
MUX
100ps
100ps
2:1
MUX
100ps
100ps
2-bit differentially
encoded at 20Gbit/s
1-bit differentially
encoded eiat
10Gbit/s
100ps
Digital description
of interferometer:
4-bit differentially
decoded hi≅diat
40Gbit/s
Figure 2.3: Simplified differential encoding/decoding scheme
2:1
MUX
4 PRBS
data
streams
with
10Gbit/s
each
2:1
MUX
2:1
MUX
100ps
Digital description
of interferometer:
4-bit differentially
decoded hiat
40Gbit/s
Figure 2.4: Realized scheme
It can be shown that the 1-bit differential decoding
hi=gi−1⊕gi,(2.6)
results again in
hi=di.(2.7)
Simplifications are once again possible if recursive differential encoding is possible at
20 Gbit/s.
Even at 10 Gbit/s, DPSK laser linewidth requirements are easily fulfilled, which allows
to simplify the matter further. If the interferometer delay is chosen equal to 4-bit durations
a more simplified scheme is possible as shown in Figure 2.3. Thus, the 4-bit differential
decoding results in
hi=ei⊕ei−4=di.(2.8)
2.1 oDPSK signal generation 11
In hereafter presented 40 Gb/s DPSK transmission experiments, differential encoding was
neither implemented nor needed because a 27−1pseudo random binary sequence (PRBS)
was transmitted. Therefore, the simplified scheme as shown in Figure 2.4 is realized in the
our laboratory.
2.1.2 Optical Phase Modulation
The complex electric field of a typical laser source used in the optical communication is
represented by: −→
E=A(t)−→
e(t) cos {ωt +ϕ(t)}(2.9)
where A(t)is the amplitude of the optical field, ωis the optical angular frequency, ϕis
the optical phase and −→
erepresents the polarization vector of the laser source, also known
as Jones-vector. These four parameter represent the four degrees of freedom that can be
exploited in the generation of modulated optical signals. Each of these parameters can be
modulated by an electrical binary baseband signal q(t):
q(t) =
∞
X
i=−∞
Ii·q(t−iTb)(2.10)
with the i-th information coefficient I∈[0,1] and the baseband pulse shape q(t)delayed
by multiples of the bit period Tb.
+1 -1 +1 -1 VPSK
a)
b)
c)
180°180°0°180°180°0°
Time, t
Time, t
Time, t
Figure 2.5: Principle of optical PSK signal modulation a) data signal b) carrier c) PSK signal
Depending on which parameter of the laser source is modulated, the modulation is
mainly differentiated as: amplitude shift keying (ASK), frequency shift keying (FSK),
phase shift keying (PSK) or polarization shift keying (PolSK).
12 Chapter 2 oDPSK Transmission System
Phase modulation is a form of data modulation scheme, where phase of the transmitted
signal is varied to convey information (Figure 2.5) but the frequency of the carrier remains
constant. The simple method, binary (D)PSK, uses only two signal phases: 0and πrad.
The digital signal is broken up time wise into individual bits (binary digits). The state of
each bit is determined according to the state of the preceding bit. It encodes 0phase shift for
a logic input and a πphase shift for a logic 0input. Thus, signal in D(PSK) representation
is given by
ϕsignal(iTb)−ϕsignal[(i−1)Tb] = {0,if q(t−iTb) = 0
π, ifq(t−iTb) = 1 (2.11)
The phase modulated optical signal could either be generated using commercially available
external Lithium Niobate-based phase modulator or by a Mach-Zehnder modulator.
Simple Phase Modulator
A phase modulator is the simplest waveguide electrooptic device where electro-optically
induced refractive index change causes a phase shift of the guided light. Figure 2.6 illus-
trates the LiNbO3-based waveguide electrooptic phase modulator.
electrical input, V
optical input optical output
Al electrodes
strip waveguide
Figure 2.6: Lithium Niobate-based Phase Modulator
Basically, phase modulator simply consist of a Ti in-diffused optical channel waveguide
placed in between the set of uniform electrodes of length, L, separated by a gap, G. The
modulating voltage waveform V, applied to the both electrodes causes electro-optically
induced refractive index change and hence the phase length variation of the channel. Ac-
tually, the refractive index of the material causes light to travel at a speed inversely pro-
portional to the refractive index of the material. Thus, if the refractive index of a material
suddenly increase, the light beam slows down and vice versa. The effective electrooptically
induced refractive index change is given by
∆n(V) = −n3
orij
2
V
GΓ,(2.12)
2.1 oDPSK signal generation 13
where nois the ordinary refractive index of LiNbO3,rij is the relevant electrooptic coeffi-
cient, an inter electrode gap G, and Γis the overlap integral between the applied electro-
static field and the optical mode [50].
Phase modulators are generally characterized by the voltage Vπwhich is defined as the
voltage required to obtain a phase shift of πrad. However, voltage–length product is more
useful to compare the performance of different phase modulators and is defined as
VπL=λG
n3
orijΓ.(2.13)
The output electrical field of the phase modulator is proportional to exp(jπV/Vπ).
Microwave contact
Optical waveguide
X-cut Z-cut
y
yz
zx
x
Figure 2.7: Waveguide based travelling-wave phase modulator using x- or z-cut LiNbO3materials
Figure 2.7 shows two waveguide based travelling-wave phase modulator structures us-
ing x-cut and z-cut LiNbO3crystal, respectively. The electric field lines are along the z-axis
in both cases. In x-cut crystal, the electrodes for the radio frequency (RF) transmission line
are located on either side of the optical waveguide where as in z-cut crystal, the electrode
for the RF transmission line are located directly on the top of the optical waveguide.
Mach-Zehnder Modulator
The operational principle of MZM are also based on the electro-optic effect. The Mach-
Zehnder waveguide structure is typically realized in LiNbO3using titanium-diffused tech-
nology. Figure 2.8 is a schematic drawing of such a modulator. At its optical input port,
there is an optical power splitter that divides the input optical power into two equal por-
tions. The divided power propagates in two separate waveguides that are often called “two
arms”.
In a MZM, at least one of the these two arms is designed as an EO waveguide, along
which the optical phase can be modulated by an applied voltage. If the optical waves are
in phase after propagating through two arms, they combine as a single mode in the output
optical combiner at the output; whereas if the optical waves are out of phase after propa-
gating both arms, they combine as a higher order spatial mode near the optical combiner,
therefore, most of the optical power is radiated into the substrate and the output intensity is
at its minimum.
14 Chapter 2 oDPSK Transmission System
electrical input, V
light in modulated light out
central electrode
Figure 2.8: X-cut Lithium Niobate-based Mach-Zehnder modulator
The optical field amplitudes at the output port of the MZM can be generally represented
by
−−→
Eout =1
√2(−→
E1ejΦ1+−→
E2ejΦ2),(2.14)
where −→
E1and −→
E2represent the optical field amplitudes in the both arms and Φ1and Φ2
represent the optical phase delays. The output optical power is
POUT =|Eout|2=1
2[|E1|2+|E2|2+ 2|E1||E2|cos(Φ1−Φ2)].(2.15)
Dividing the POUT by the input optical power PIN = (|E1|2+|E2|2)of the MZM and after
some parameter transformations, the optical intensity transfer function for the MZM can
be written in the form of
TMZM =1
2[1 + bcos(Φ1−Φ2)],(2.16)
where b= 2|E1||E2|/(|E1|2+|E2|2)is an optical imbalance factor between the two arms.
b= 1 for ideally balanced design. The phase difference (Φ1−Φ2)consist of two parts;
one is the path difference Φ0at zero applied voltage and the other is the phase difference
∆Φ due to applied voltage.
When only one arm is modulated the phase difference becomes
∆Φ = 2π
λ∆nLΓ,(2.17)
where ∆nis the electrooptically induced index change, Γis the overlap integral factor,
λis the wavelength, Lis length of the modulator electrode. If both arms are modulated
in push-pull mode, which means that the phase changes in the two arms are opposite, the
overall phase change ∆Φ is simply doubled.
2.1 oDPSK signal generation 15
If the modulation is based on the electrooptic effect, then
∆n=1
2n3
orij
V
G,(2.18)
where nois the optical index at the zero applied voltage, rij is the relevant electrooptic
coefficient, Vis the applied voltage and Gis the inter electrode gap. Combining (2.17) and
(2.18), one gets
∆Φ = π
λn3
orij
V
GLΓ = πV
Vπ
(2.19)
and
Vπ=λG
n3
orijΓL.(2.20)
Vπis a very important parameter for MZM. It is voltage required to induce a phase differ-
ence of πrad.
Figure 2.9: Three different structures for Mach-Zehnder modulator using x- or z-cut LiNbO3
Figure 2.9 shows the different electrode structures used in LiNbO3-based MZM’s de-
pending on the crystal cut and the propagation direction. In order to exploit the highest
electrooptic coefficient, the strongest component of the applied electrical field must be
aligned with the z-axis of the crystal. For x-cut LiNbO3crystal, an electrical field along
the z-directions means a horizontal electrical field whereas for z-cut LiNbO3crystal, an
electrical field along the z-directions means a vertical electrical field [51–53].
16 Chapter 2 oDPSK Transmission System
MZM Versus Phase Modulator
For DPSK signal generation, the MZM is biased at its transmission minimum and needs a
voltage swing of 2Vπ. MZM has highly accurate phase modulation at the expense of resid-
ual intensity modulation. This results in the intensity dips [8] with the widths depending
on the drive signal. Since DPSK encodes the information in the optical phase rather than
in the intensity, these dips are absolutely of no importance.
Chromatic dispersion (CD) tolerance (Section 2.3.1) of DPSK signals generated either
using a Mach-Zehnder modulator or a phase modulator at the data rate of 40 Gbit/s is
evaluated by simulation studies at the chromatic dispersion of 0, 34, and 68 ps/nm. Fig-
ure 2.10 shows the simulation results [54]. The DPSK signal generated with Mach-Zehnder
modulator exhibits better CD tolerance with respect to the DPSK signal generated using
the phase modulator. Therefore, our DPSK transmission setup uses a dual drive MZM at
40 Gbit/s [55].
Mach-Zehnder
modulator
Phase
modulator
34 ps/nm0 ps/nm 68 ps/nm
Figure 2.10: Chromatic dispersion tolerance of DPSK using either a Mach-Zehnder modulator or a phase
modulator at the data rate of 40 Gbit/s and chromatic dispersion of 0, 34, and 68 ps/nm
2.1.3 40 Gbit/s (CS)RZ-DPSK transmitter
Figure 2.11 shows the in-house developed 40 Gbit/s RZ-DPSK transmitter. The electri-
cal part of the transmitter basically employs 16:1 Infineon multiplexer that processes 16
2.5 Gbit/s signals to generate the 40 Gbit/s signal. Tx also uses SHF modulator drivers for
the Agere’s dual drive data modulator and another dual drive modulator as a pulse carver,
driven at half of the clock rate to generate either CSRZ (66% duty cycle) or RZ (33% duty
cycle) signals. The details of the Tx are given in the next subsections.
2.1 oDPSK signal generation 17
16 data buffers
laser MUX
pulse
modulation
NRZ DPSK
modulation
VCO
Figure 2.11: 40 Gbit/s CSRZ-DPSK transmitter
Data buffer board
16:1 multiplexer multiplexes 16 data streams at 2.5 Gbit/s to generate 40 Gbit/s 27−1PRBS
data. Therefore, this multiplexer requires a data buffer board that will generate 16 outputs
from the Pulse Pattern Generator’s (PPG) data output. Data buffer board (Figure 2.12) is
designed and developed in-house to provide 16 data output streams which are mutually
delayed by multiples of 8 bits.
CLK and CLK
from PPG
16 data outputs at 2.5Gb/s
Clock amplifiers
Delay
lines
2.5Gb/s input
signal from PPG
Figure 2.12: Photograph of the data buffer board
18 Chapter 2 oDPSK Transmission System
It is realized on PCB board having 6 layers using the four OnSemi MC100EP131 Quad
Master-slaved D flip-flops with common set and separate resets. 27−1PRBS data is
differentially clocked into the flip flops at 2.5 Gbit/s using the PPG clock outputs (CLK
and CLK). Differential clock signals are amplified before going to the flip-flop inputs
using the monolithic amplifiers ERA-1 having a typical gain of 10 dB at 3 GHz.
Multiplexer
The multiplexing is done in a SiGe-based Infineon Multiplexer (MUX) which multiplexes
16 2.5 Gbit/s inputs to a single 40 Gbit/s output. This 16:1 MUX has the conventional
tree-type architecture which has built in 16:8, 8:4, 4:2, and 2:1 internal multiplexing units.
The input 2.5 Gbits/s data streams are mutually delayed by multiples of 8 bits so that the
resulting 40 Gbit/s pattern is also a PRBS pattern.
Delay Flip–Flop
Waveform reshaping and retiming function is done at 40 Gbit/s using the NEL (CI0085B)
delay flip-flop (D-FF). It is based on 0.1µm InP-based HEMT process. It operates up to
the data rates of 43 Gbit/s and beyond. For a single ended clock and data input, it provides
0.9 Vpp differential output signals which are directly connected to the two inputs of the
SHF modulator drivers that drives the dual drive data modulator.
Modulator Driver
SHF 806E is a modulator driver amplifier having three stage design. Driver is a monolithic
microwave integrated circuit hermetically packaged to achieve ultra wide bandwidth and
low noise performance. It has gain of ∼26 dB and bandwidth >38 GHz. Total power
consumption of the driver amplifier is about ∼6W and it delivers an output signal having
8 Vpp amplitude into 50 Ω load.
40 Gbit/s LiNbO3Electro-Optic Modulator
Tx uses the distributed feedback (DFB) laser at 192.5 THz. The laser output is connected
to the input of the dual drive data modulator. The use of dual drive technology inherently
offers the capability to adjust the modulator chirp for ASK mode of operation and chip-free
operation for DPSK mode of operation. Dual drive modulators are able to operate in the
wavelength range of 1525-1620 nm. They have a maximum optical insertion loss of 6 dB,
minimum DC and RF extinction ratios in the order of 20 dB and 14 dB, respectively and
minimum bandwidth of 30 GHz. Maximum drive voltage at 1 GHz is about 3 V per side.
2.1 oDPSK signal generation 19
Generation of optical NRZ-DPSK signal
In house developed 40 Gbit/s DPSK transmitter is shown in Figure 2.13. When using a
MZM for phase modulation, the modulator is biased at its point of minimum transmission
and is driven at twice the voltage swing required for ASK mode of operation. If z-cut MZM
is used then it is driven in push-pull mode of operation to minimize the chirp, whereas an
x-cut modulator requires only a single electrical drive.
NRZ Data
NRZ Data
Clock
Clock
RZ-DPSKNRZ-DPSK
„0“ line „0“ line
Figure 2.13: DPSK Transmitter using the MZMs
The method of generating the optical binary phase modulation by using a dual–drive
MZM is shown in Figure 2.14. Since the phase of the optical field changes its sign upon
transitioning through a minimum in the MZMs power transmission curve, two neighbor-
ing intensity maxima have opposite optical phases, and a near-perfect 180◦phase shift is
obtained, independent of the drive voltage swing. The benefit of highly accurate phase
modulation comes at the expense of some residual intensity modulation at the transition
of two bits, with the width of the resulting intensity dips depending on the drive signal’s
bandwidth and voltage. However, DPSK encodes information in the optical phases rather
than in the intensity, these dips are of no importance, especially for RZ-DPSK.
A digital “1” is represented by a πphase change between the consecutive data bits in
the optical carrier, while there is no phase change between the consecutive data bits in the
optical carrier for a digital “0”. For NRZ-DPSK signal optical power is constant, however
the optical field shifts between “1” and “-1” , that what it differs from ASK, where the
optical field shifts between “1” and “0”.
To improve system tolerance to nonlinear distortion and to achieve a longer transmis-
sion distance (higher sensitivity), instead of NRZ-DPSK, return-to-zero DPSK (RZ-DPSK)
is often used. But, to generate the RZ-DPSK optical signal, one more intensity modulator
needs to be used as pulse carver.
20 Chapter 2 oDPSK Transmission System
MZ Transmission
Optical Power Drive Voltage
Optical Field
Time
Optical Power
Im{E}
DPSK Constellation
Points
Re{E}
Intensity Dips
Intensity Dips Time
π0πππ0
DPSK
DPSK Drive
Signal
Figure 2.14: DPSK signal generation
Generation of optical (CS)RZ-DPSK Signal
(CS)RZ-DPSK signal is generated in our transmitter using another dual drive modulator
driven with a sinusoidal signal at half of the clock rate. Figure 2.15.a shows the method
of generating the 66% and 33% RZ signals by biasing the modulator at transmission min-
imum and maximum respectively, and driving the modulator with a sinusoidal signal with
a maximum drive voltage swing of 2Vπfor both cases. Figure 2.15.b shows the resulting
intensity pulses for both the 33% and 66% (CS)RZ pulses.
Drive Voltage
MZI Transmission
66% RZ
33% RZ
0 1 2 3 4 5 6 7 8 9 10
33% RZ 66% RZ
Tbit Tbit
Intensity
Intensity
a) b)
Figure 2.15: a)Transmission characteristic of MZM b)Optical intensity generated signals
2.2 oDPSK signal detection 21
2.1.4 Experimental Results
To judge the quality of the modulation, NRZ- and RZ-DPSK optical signals are first gen-
erated using the developed hardware and then received using two high-speed photodiodes.
Output of the two photodiodes are directly connected to a 50 GHz oscilloscope. The os-
cilloscope was triggered with the transmitter clock at 2.5 GHz to view the eye diagrams
at 40 Gb/s. The exemplary NRZ- and RZ-DPSK eye diagrams at the transmitter side are
shown in Figure 2.13.
Since the pulse width of the (CS)RZ-DPSK signal is narrower than that of the NRZ-
DPSK signal, the (CS)RZ-DPSK pulse has higher peak power than the NRZ-DPSK for a
given average power. Thus, the eye opening of the (CS)RZ-DPSK signal format is wider
than that of the NRZ-DPSK (inset Figure 2.13), resulting in better receiver sensitivity than
the NRZ-DPSK for a given average power [23]. This implies that for a required receiver
sensitivity, the transmitted power can be lowered by employing the (CS)RZ-DPSK signal
format rather than the NRZ-DPSK. The better receiver sensitivity in the case of the (CS)RZ
signal also suggests that the transmission distance can be increased compared with the
NRZ-DPSK signal for the same transmitted power.
2.2 oDPSK signal detection
A typical optically pre-amplified balanced DPSK receiver is shown in Figure 2.16. The
optical signal is first amplified using a pre-amplifier. Then it passed through an optical
band pass filter in order to improve the optical signal-to-noise ratio before it enters the
Mach-Zehnder delay-interferometer (MZDI), whose differential delay is set equal to the
bit period. This optical signal preprocessing is necessary in direct-detection receivers to
accomplish demodulation, since the photodetection process is insensitive to the optical
phase; a detector only converts optical intensity modulation into an electrical signal.
optical
input
optical
amplifier
optical
bandpass
filter Mach-Zehnder
interferometer differential photoreceiver with
lowpass characteristic
electrical
output
Figure 2.16: DSPK optical receiver
22 Chapter 2 oDPSK Transmission System
In direct-detection DPSK receiver, the MZDI lets two adjacent bits interfere with each
other at its output port. This interference leads to the presence (absence) of power at
a MZDI output port if two adjacent bit interfere constructively (destructively) with each
other. Thus, the preceding bit in a DPSK encoded bit stream acts as the phase reference
for demodulating the current bit. Two MZDI output ports generally carry identical, but
logically inverted data streams under DPSK modulation. MZDI’s can be fiber based or else
could be implemented as a planer lightwave circuit (PLC) technology.
The DPSK using balanced detection, has the most obvious benefit. It exhibits 3 dB
lower OSNR required to reach a given BER compared to conventional ASK. At a BER of
10−9, the quantum limit for an optically preamplified ASK receiver is 38 photons/bit [56],
however only 20 photons/bit are needed for the optically preamplified balance DPSK re-
ceiver [13]. A receiver sensitivity of 36.2 dBm (45 photons/bit) was reported for a 42.7 Gb/s
optically preamplified return-to-zero DPSK signals [11]. However, the reported record sen-
sitivity is 38 photons/bit at 42.7 Gb/s once again using RZ-DPSK signals [12].
2.2.1 Mach-Zehnder Interferometer Modelling
A Mach-Zehnder delay interferometer (MZDI) consists of a input 2x2 coupler, an output
2x2 coupler and two waveguide branches in between them with unequal optical path length
difference corresponding to the integer multiples of bit duration (Figure 2.17). Thus, the
MZDI’s power transfer matrix is given by
SMZDI = [Scoupler][Sbranches][Scoupler].(2.21)
The upper arm of the Mach-Zehnder interferometer is a direct connection and the lower
arm contains the delay including the phase adjustment.
Delayed arm
E1
E2
E
Figure 2.17: Simplified Interferometer model with delay T and retardation R in the lower branch
As the both branches are summarized by
[Sbranches] = 1 0
0e−(jωT+ϕ)(2.22)
2.2 oDPSK signal detection 23
the equation (2.21) becomes
SMZDI =1
√21−j
−j11 0
0e−(jωT+ϕ)1
√21−j
−j1(2.23)
If the electrical field at the MZI input is −→
E(t), both output arms of the splitter in the MZI
carry electric fields −→
E(t)/√2. Therefore the output fields at the the two photodiodes are
−→
E1(t) = 1
2−→
E(t) + −→
E(t−T)(2.24)
and −→
E2(t) = 1
2−j−→
E(t) + j−→
E(t−T)(2.25)
As the two adjacent bits interfere with each other at the MZDI output ports, the optical
output of the Mach-Zehnder delay interferometer is either a constructive or a destructive
interference depending on the relative phase difference between −→
E(t)and −→
E(t−T). The
signals measured by the two photodiodes are
|−→
E1(t)|2="−→
E(t) + −→
E(t−T)
2#2
(2.26)
and
|−→
E2(t)|2="−→
E(t)−−→
E(t−T)
2#2
(2.27)
For the difference of intensities at the two photodiode outputs, one can write
|−→
E1(t)|2−|−→
E2(t)|2=Re(−→
E+(t)·−→
E(t−T)).(2.28)
Polarization matching is achieved by a polarization-independent design of the MZDI.
2.2.2 40 Gbit/s RZ-DPSK receiver
In the receiver, after passing an optical preamplifier (EDFA) and an optical BPF filter (Fig-
ure 2.16), the optical signal enters a commercial Mach-Zehnder interferometer having four
bit delay (NEL). This allows us to differentially encode the data at 10 Gbit/s in the transmit-
ter (Section 2.1.1). The two outputs from the Mach-Zehnder interferometer are connected
to two high-speed photodiodes from u2t.
24 Chapter 2 oDPSK Transmission System
A variable electrical delay line was used to adjust the path length difference between the
two photodiode outputs. The two photodiode outputs are directly connected to the differ-
ential inputs of a Infineon demultiplexer having standard clock and data recovery circuits
(CDR). From the received NRZ data stream, either at 40 or at 43 Gbit/s, CDR recovers
the clock and data. Demultiplexer demultiplexes the 40 Gbit/s data to 16×2.5 Gbit/s data
streams. Additional details about the components used in the DPSK receiver are given in
the following subsections.
Mach-Zehnder Interferometer
NEL Mach-Zehnder delay interferometer(MZDI) is planar lightwave circuit (PLC) based
on silica on silicon waveguide technology. It is stable against mechanical vibration. The
MZDI is temperature stabilized at 45◦C using a proportional integral (PI) temperature con-
troller. The free spectral range of the interferometer is 10 GHz. Its phase tuning speed is
in the range of 2-3 ms and the tuning range is between 0−2πrad. Its insertion loss is
≤3.5dB and extinction ratio is >17 dB. The phase tuning is accomplished by using the
differential micro-heaters with total constant power.
Photo Receiver
The u2t (XPDV202R) photodiodes have bandwidths >50 GHz and the responsivities of
0.68 A/W. The maximmum dark current is 500 nA. Photodiodes are equipped with an
internal 50 Ωresistor between RF-signal and ground, so they can be biased via the bias-pin
and ground-pin regardless of a measurement unit being connected. Typically, a bias voltage
in the order of 2 V is needed. The maximum allowed optical input power is 20 mW. The
average photocurrent on each detector was about 2.5 mA which corresponds to average
optical power of ≈5dBm.
Lock-In Stabilization Scheme
To make the delay interferometer more stable and more robust against the existing small
polarization dependent phase shift, the lock-in stabilization scheme was first proposed and
later implemented by R. No`
e, using the existing lock-in amplifier board. Parts of the pho-
todiode output signals are tapped off using pick-off T’s and sent through the 40 Gbit/s
Infineon differential amplifiers for subsequent AC power detection. Detected RF power is
at its maximum when the interferometer phase difference is set to 0◦. The DC voltage that
controls the differential micro-heaters with total constant power was therefore added with
a small signal 400 Hz tone to thermally modulate the interferometer phase difference for
subsequent lock-in stabilization. The detected RF power was synchronously detected using
a lock-in amplifier which utilizes an AD734 as an analog multiplier. The reference signal
was fed into the second input of the multiplier. An additional phase shifting network was
used in the reference path for obtaining the best sensitivity from the lock-in amplifier.
2.2 oDPSK signal detection 25
The multiplier’s output signal was low pass filtered using a proportional integrator
whose output finally controls the interferometer phase difference. Due to lock-in stabi-
lization scheme (Figure 2.18), when the interferometer phase difference is set optimally by
the controller, the eye diagram is open and the 400 Hz tone is absent. An opposite thing
happens, when the phase difference is set incorrectly, so that the eye diagram is closed and
the 400 Hz tone is present.
Phase shifter
Output Amplifier
Multiplier
Infineon differential
40 Gbit/s amplifiers
RF diode
Pick-off T´s
MZI
100 ps
clock and data
recovery
MZI lock-in stabilization
Input amplifier
Differential Microheaters with
total costant power circuit
Summer
Figure 2.18: Block diagram of the lock-in amplifier’s scheme
2.2.3 Measurement results
The measured results show that difficulties in implementing stable delay interferometers
[57] have been overcome using the proposed lock-in stabilization scheme. This 400 Hz
lock-in stabilization scheme for the interferometer phase essentially eliminates the impact
of a small polarization-dependence of the interferometer phase shift.
-80
-60
-40
-20
0
20
1554 1556 1558 1560
Wavelenght [nm]
Power [dBm]
-80
-60
-40
-20
0
20
1554 1556 1558 1560
Waveght [nm]
Power [dBm]
Figure 2.19: Optical spectrum at the constructive port (left) and destructive port(right)
26 Chapter 2 oDPSK Transmission System
The interferometer phase difference could be continuously fine-tuned in order to match
the laser center frequency. As mentioned above, the two MZDI’s output ports carry in-
verted data streams. Figure 2.19 shows the optical spectrum recorded at constructive and
destructive port of the MZDI in a direct-detection DPSK receiver. Since both ports carry the
full (only inverted) information, they can be detected either using a so called “single-ended
detection” or using “balanced detection”.
Figure 2.20 shows the experimentally measured BER curves for balanced detection
as well as single-ended detection of (CS)RZ-DPSK signals using in-house developed op-
tically pre-amplified receiver. The corresponding back-to back receiver sensitivities are
−33.5dBm and -27.5dBm.
-40 -38 -36 -34 -32 -30 -28 -26
10-11
1x10-7
1x10-3
RZ-DPSK
BER
Sensitivity [dBm]
Single-ended
RZ-DPSK
Figure 2.20: BER vs. power at optical preamplifier input for different CS-RZ DPSK modulation format
Figure 2.21 shows the measured back-to-back eye diagrams for NRZ-DPSK as well as
for CSRZ-DPSK signals.
Figure 2.21: 40 Gbit/s eye diagrams back-to-back for NRZ-DPSK (left) and (CS)RZ-DPSK (right)
2.3 Signed On Line Chromatic Dispersion Detection 27
2.3 Signed On Line Chromatic Dispersion Detection
Tunable chromatic dispersion (CD) compensation is often needed in long-haul and dynam-
ically routed transmission links, especially at 40 Gbit/s and beyond. Among many CD
detection schemes, a synchronous arrival time detection was implemented for intensity-
modulated systems [58]. In this scheme arrival time modulations caused by a small fre-
quency modulation in the presence of CD are synchronously detected in the clock recovery
phase locked loop (PLL). In this work, it is demonstrated that this signed online CD de-
tection scheme also works for the DPSK modulation format, even with an interferometer
having 100 ps delay [59].
2.3.1 Chromatic Dispersion in Single Mode Fibers
The main advantage of single-mode fibers is that intermodal dispersion is absent simply
because the energy of the injected pulse is transported by the single optical mode. How-
ever, the pulse broadening does not disappear altogether. The group velocity associated
with the fundamental mode is frequency dependent because of chromatic dispersion. As
a result different spectral component of the pulse travels slightly different group veloci-
ties, a phenomenon referred to as the Group Velocity Dispersion or simply the linear fiber
dispersion.
The concept of chromatic dispersion (CD) or group velocity dispersion can be under-
stood as follows: the complex field transfer function of an optical fiber of length Lis
H(ω) = e−jβ(ω)L(2.29)
where β(= nk0=nω/c)is the propagation constant, nis the modal index and ωis the
optical angular frequency.
It’s phase can be approximated, by neglecting the fiber attenuation, by the truncated
Taylor series expansion as
ϕ=−β(ω)·L= (β+ (ω−ω0)β0+1
2(ω−ω0)2β00)·L. (2.30)
At ω0the propagation constant β(ω)assumes the value β, its first and second derivative
with respect to ωare β0and β00, respectively. The group delay
τg=−ϕ0= (β0+ (ω−ω0)β00)·L(2.31)
is a linear function of ω. It’s derivative with respect to wavelength λand length Lis the
chromatic dispersion coefficient
D=d2τg
dλ ·dL =−2πc
λ2β00 (2.32)
The β00 in (2.32) is nothing but the derivative of V−1
g(= dβ/dω)with respect to ω,
where the group velocity Vgis given by Vg=c/ng. The group index ng=n+ω(dn/dω).
28 Chapter 2 oDPSK Transmission System
Therefore, the wavelength dependence of Das shown in (2.32) is in fact governed by the
frequency dependence of the modal index nand can be written as [60]
D=−2πc
λ2
d
dω(1
Vg
) = −2π
λ2(2dn
dω +ωd2n
dω2).(2.33)
1.1 1.2 1.3 1.4 1.5 1.6 1.7
Wavelenght [µm]
Dispersion
0
Material Dispersion
Chromatic Dispersion
Waveguide Dispersion
Zero Dispersion
Wavelengh at 1.31 µm
Figure 2.22: Total dispersion Dand relative contributions of material dispersion Dmand waveguide disper-
sion Dwfor a conventional single mode fiber
Generally, Dcan be written as the sum of the two terms D=Dm+Dw, where Dmand
Dware material and the waveguide dispersions, respectively. Material dispersion occurs
because the refractive index of silica, the material used for fiber fabrication changes with
the optical frequency ω, while the waveguide dispersion arises because the way in which
the waves match the boundary conditions at the core-cladding interface depends on their
frequency. As a result, their propagation phase velocity is a function of frequency, inde-
pendent of any material effects. Another way of understanding this is to recognize that, in
monomode fibers, a significant fraction of the optical power propagates in the cladding. As
the frequency varies, so the propagation of the power travelling in the cladding changes,
and so the average refractive index experienced by the waves also changes. Figure 2.22
shows Dm,Dwand their sum, D=Dm+Dwfor standard single mode fiber.
2.3.2 Measurement Setup for Chromatic Dispersion Detection
Adjustable drop-in CD compensator requires a signed online CD detection. For this pur-
pose the DFB transmitter laser is frequency-modulated at 5 MHz with a 224 MHz (rms)
frequency deviation, and a parasitic 1.2% (rms) amplitude modulation. Figure 2.23 shows
the 40 Gbit/s in-house developed CSRZ-DPSK transmission setup with signed online chro-
matic dispersion detection.
2.3 Signed On Line Chromatic Dispersion Detection 29
In the presence of chromatic dispersion (CD), the frequency modulation (FM) causes
a small arrival time modulation which is indicated by the clock phase error signal. This
arrival time signal is synchronously detected using a multiplier and averaging circuit. A
low-frequency monitor photodiode with bandpass filter at 5 MHz detects the amplitude
modulated (AM) reference signal for a 5 MHz lock-in detection of the clock phase error
signal coming from the clock recovery phase locked loop (PLL).
5MHz
FM&AM
in
20GHz
in
DFB
laser fiber
with CD
CSRZ
40Gbit/s
data in
DPSK
data
out
clock and
data
recovery
BPF
BPFAVG
CD
phase trimmer
& limiter
clock signal
PI
multiplier
VCO
AM
reference
100ps
MZI lock-in
clock phase
error signal
(arrival time)
Figure 2.23: Experimental 40 Gbit/s CSRZ-DPSK setup for chromatic dispersion detection
2.3.3 Experimental results
The function of the chromatic dispersion detection is verified by inserting various fiber
pieces. Figure 2.24 shows the CD readout as a function of true CD in the range −91 ps/nm
. . . +147 ps/nm. The readout is fairly linear in the range where the eye diagram is fairly
open. The sign of the CD is faithfully returned even when the eye diagram is closed (inset
Figure 2.24) as long as the clock phase detector works, the PLL locks, and there is a high-
enough percentage of correct data decisions. Since the sign of the CD is preserved, the CD
error signal could be directly used to control an adaptive CD compensator via an integral
controller.
30 Chapter 2 oDPSK Transmission System
CD [ps/nm]
CD [a.u.]
-100 -50 0 50 100
-2
-1
0
1
2
150
3
Figure 2.24: Chromatic dispersion detection readout vs. actual dispersion. Inset: eye diagram resulting from
interferometer output signal difference
The readout noise (rms) at zero CD (Figure 2.25), ranges from 4 ps/nm to <100 fs/nm
for measurement intervals between 38 µs and 157 ms. The 27−1PRBS yielding slightly
better results than a 223 −1PRBS.
10-410-310-210-1
0.01
1
10-5
σCD[ps/nm]
0.1
10
1
223-1PRBS
27-1 PRBS
CSRZ DPSK
Time [s]
Figure 2.25: Standard deviation versus measurement interval, at zero actual dispersion
The eye diagrams back to back at each photodiode and their difference are shown in
Figure 2.26 (left). The eye diagrams after 91 km of transmission are also shown in Fig-
ure 2.26 (right).
The Q factor is 24 dB for 17 ps CSRZ pulses. For 8 ps RZ pulses were also tried out and
yielded a Q>28 dB. The CSRZ-DPSK signal was also transmitted over 58 km of SSMF,
33 km of DSF, and some DCF. The Q factor after transmission was always >22 dB.
2.4 Chromatic Dispersion Compensation 31
Figure 2.26: CSRZ-DPSK eye diagrams at interferometer outputs (top), and difference signal (bottom) back
to back (left) after transmission over the 91km (right)
Compared to the recently reported method [61], the employed CD detection scheme [58]
needs a much smaller frequency modulation. This means reduced parasitic amplitude mod-
ulation and is of course advantageous in densely packed WDM environments. Thus, a small
FM was applied to the transmitter DFB laser allows us to measure chromatic dispersion on-
line in sub-ms intervals, including its sign. Therefore, the required frequency deviation is
so small that this scheme can be applied to DPSK modulation format even with an interfer-
ometer having 100 ps delay.
2.4 Chromatic Dispersion Compensation
At 40 Gbit/s, chromatic dispersion is the main limiting factor, as the system tolerance is
reduced to 1/16 of that at 10 Gbit/s [62]. Temperature changes can lead to variations in
dispersion that may be significant enough to degrade system performance. Therefore an
accurate, tunable CD compensation is often required. Various types of integrated opti-
cal dispersion compensators [63–67] have been demonstrated but, the fiber Bragg grating-
based dispersion compensators exhibit the largest dispersion range and lowest insertion loss
with an associated tunability.
Fiber Bragg Grating
The significant discovery of photosensitivity in optical fibres led to the development of a
new class of in-fibre components, called the fibre Bragg gratings (FBGs). In its simplest
form a fiber Bragg grating consists of a periodic modulation of the index of refraction
in the core of a single-mode optical fiber. These types of uniform fiber gratings, where
the phase fronts are perpendicular to the fiber’s longitudinal axis and with grating planes
having constant period, are considered as the fundamental building blocks for most of the
32 Chapter 2 oDPSK Transmission System
fiber Bragg grating structures. Generally, if the light propagating in the fiber core having
the above mentioned refractive index modulation satisfies the well known Braggs condition
then and only then it is strongly coherently reflected back.
The Bragg grating condition is simply the requirement that satisfies both energy and
momentum conservation principles. Energy conservation (¯hωi= ¯hωr) requires that the
frequency of the incident radiation and the reflected radiation is the same [68]. Momen-
tum conservation requires that the incident vector ki, plus the grating vector, Kequal the
wavevector of the scattered radiation kr. This is simply stated as
ki+K=kr,(2.34)
where the grating wavevector, K, has a direction normal to the grating planes with a mag-
nitude 2π/Λ(Λis grating spacing shown in Figure 2.27).
λbroad-λb
λbroad λb
Λ
Ki Kr
K
Bragg Grating
Figure 2.27: Illustration of a uniform grating with constant amplitude of refractive index modulation and
grating period
The diffracted wavevactor is equal in magnitude, but opposite in direction, to the inci-
dent wavevector. Thereby, for the momentum conservation condition stands
22πneff
λB=2π
Λ(2.35)
which simplifies to the first-order Bragg condition
λB= 2neffΛ(2.36)
where the Bragg grating wavelength, λB, is the free-space center wavelength of the input
light that will be reflected from the Bragg grating, and neff is the effective refractive index
of the fiber core at the free space center wavelength.
2.4 Chromatic Dispersion Compensation 33
Consider a uniform Bragg grating formed within the core of an optical fiber with an
average refractive index n0, the index of refractive profile can be expressed as
n(z) = n0+1
2∆npp(z) cos(2π
Λz+φ(z)),(2.37)
where ∆npp(z)is the gratings peak to peak refractive index modulation amplitude (typical
values 10−5to 10−3), and φ(z)is the grating phase. Lis the grating length, and zis the
distance along the fiber longitudinal axis.
Dispersive Fiber
Dispersed Pulse
Circulator
λred
λblue
In Out
Figure 2.28: Principle of FBG CD compensator with circulator
Devices with the fiber Bragg grating are often used in conjunction with optical circu-
lator. The light normally enters the circulator input port (1) and appears at the second port
(2), which is connected to the FBG. Then, the light travels through the FBG and reflects
back to port 2. The light that enters the port 2 is routed to output port (3) (Figure 2.28).
The fiber Bragg grating introduces the negative dispersion to clean the spectrum of the the
dispersed signal. Dispersion compensators based on optical FBGs are now commercially
becoming the promising candidates for tunable dispersion compensation [69,70]. The ad-
vantages of FBGs are large nonlinear tolerance and lower device insertion loss. The main
drawback associated with the FBG-based dispersion compensators is distortions arising
from the residual amplitude and phase ripples of the Bragg grating as well as the increased
system complexity due to the fact that such compensators need high resolution temperature
controllers for their operation.
2.4.1 Adaptive Tunable CD Compensation
Tunable chromatic dispersion (CD) compensation is needed in long haul and dynamically
routed transmission links, especially at 40 Gbit/s. Among many CD detection schemes,
synchronous arrival time detection with a sensitivity of at least 200 attoseconds [58] is the
most promising option because the scheme has an extremely low incremental cost, provides
the sign of CD and is usable for various modulation formats [59].
34 Chapter 2 oDPSK Transmission System
The tolerance to residual CD with respect to in-line CD compensation ratio for vari-
ous modulation formats including NRZ-ASK, CSRZ-ASK, NRZ-DPSK and CSRZ-DPSK
was evaluated numerically in [71] at 43 Gbit/s. But in [23], it is for the first time re-
ported on automatic chromatic dispersion compensation for all these modulation formats
in a 40 Gbit/s transmission experiment, using a thermally tunable dispersion compensator.
The fully automatic residual chromatic dispersion compensation is reported for a fiber link
with fibers up to 263 km in length for NRZ-DPSK as well as CSRZ-DPSK modulation
formats at 40 Gbit/s, using synchronous arrival time detection and a thermally tunable dis-
persion compensator.
Transmission setup
Figure 2.29 shows 40 Gbit/s DPSK transmission setup for adaptive tunable CD compen-
sation. A DFB laser at 192.5 THz (1557.366 nm) is modulated with a 5 MHz sinusoidal
source to provide 1.8% (rms) power modulation and 336 MHz (rms) frequency modula-
tion. A 40 Gbit/s 27−1PRBS is impressed on the optical carrier to generate NRZ-DPSK
or CSRZ-DPSK signal for transmission (Section 2.1.3).
100 ps
CD
error
DCFDCF
DFB
laser DPSK
5 MHz
FM&AM 40 Gbit/s
data
out
clock
anddata
recovery
PI
clock
VCO
clock
phase
error
signal
tunableCD
compensator 5 MHz
BPF
controller
5 MHz
BPF
locking status
90
km
84
km
89
km
20 GHz
CSRZ
Figure 2.29: CDC Setup for 40 Gbit/s DPSK transmission
This signal is transmitted over three fiber spans with a total length of 263 km. The
spans were mixed from 170 km of SSMF, 60 km of NZDSF, and 33 km of DSF. DCF with
a total dispersion of -2713 ps/nm was inserted between first and second stages of the two
inline EDFA’s.
2.4 Chromatic Dispersion Compensation 35
Figure 2.30: Photograph of the TeraXion thermally tunable dispersion compensator
At the receiver end, there is an 980 nm optical preamplifier followed by a 40 channel,
flat top wavelength division Optun de-multiplexer which is being used as a narrow band
optical band pass filter. The TeraXion tunable dispersion compensator (TDC) (Figure 2.30)
is inserted just before the receiver using a three-port optical circulator. This single-channel
TDC [69] at 192.5 THz is based on thermally tunable chirped FBG. The dispersion tuning
range of device is between -300 and -700 ps/nm. Insertion loss excluding circulator loss is
less than 1.5 dB. Figure 2.31 shows the group delay versus wavelength in tunable chromatic
dispersion compensator for various dispersion settings.
-900
-800
-700
-600
-500
-400
-300
-200
1556,85 1557,05 1557,25 1557,45 1557,65 1557,85
Wavelength [nm]
Group Delay [ps]
Channel Bandwidth = 85 GHz
-700
-600
-500
-400
-350
-300
D2[ps/nm]
Figure 2.31: Group delay versus wavelength in tunable CD compensator for dispersion settings
An optical tap and a low frequency power monitor photodiode is used to recover the
power modulation whose output is being used as a reference signal for arrival time detection
followed the dispersion compensator. DPSK signals are decoded using a Mach-Zehnder
interferometer having a 100 ps delay. Both interferometer outputs are connected to high-
speed photodiodes, which in turn are connected to the differential inputs of an Infineon
clock and data recovery circuit with 1:16 DEMUX.
36 Chapter 2 oDPSK Transmission System
BERs in even and odd DEMUX channels are about the same. For ASK operation, the
interferometer and one photodiode is left out. In the presence of CD, the FM causes small
arrival time modulation, which is indicated by the clock phase error signal. The CD error
signal is directly proportional to the residual CD including its sign.
Experimental Results
In order to keep the CD readout independent of optical input power fluctuations, the de-
tected photocurrent is stabilized by feedback loop that controls the pump current of the last
EDFA. Figure 2.32 shows OSNRs in dB/0.1nm, which all result in BER = 10−9. They are
given as a function of a manually adjusted compensator CD with a fiber link of 258 km in
place.
-480 -440 -400 -360 -320
24
26
28
30
32
34
CSRZ-DPSK
NRZ-DPSK
CD [ps/nm]
OSNR at BER=10-9[dB/0.1 nm]
Figure 2.32: OSNR needed for BER = 10−9versus compensator CD
Automatic CD compensation is carried out by a Digital Signal Processor (DSP) in
the following way: at first, the dispersion compensator is thermally scanned through its
−300 ps/nm to −700 ps/nm CD tuning range. Then it is set into the middle of that region
where the clock recovery PLL locks successfully. Finally, an integrator controls the value
of the CD. The integrator input is driven by the CD error signal. Integration stalls when the
CD error signal vanishes and indicates zero residual CD. Temporal variations of CD are
automatically tracked. The electrical heating/cooling power required to control and tune
the compensator is 10 W. A thermal scan takes 10 minutes, and the control time constant is
about 45 s, but control speed was not optimized.
The back-to-back Q values for NRZ-DPSK and CSRZ-DPSK are 25.8and 29.5 dB, re-
spectively. The corresponding back-to-back receiver sensitivities of −26.8, and −32.1dBm
are equivalent to OSNRs of 29.6, and 23.8dB/0.1nm, respectively.
2.4 Chromatic Dispersion Compensation 37
18 20 22 24 26 28 30 32
10-11
10-9
10-7
10-5
10-3
4
3
2
1
4
3
2
1
1 back-to-back
2 253 km
3 258 km
4 263 km
CSRZ-DPSK
NRZ-DPSK
BER
OSNR [dB/0.1 nm]
Figure 2.33: BER versus OSNR. The OSNR is varied by an attenuator.
Figure 2.34: 40 Gbit/s eye diagrams back-to-back (top) and after 263 km transmission (bottom), for NRZ-
DPSK and CSRZ-DPSK (from left to right)
Figure 2.33 shows BER vs. OSNR. With the 263 km fiber link, the Q factors are re-
duced to 19.1and 20.4dB, respectively, and they stay essentially unchanged when the
tunable dispersion compensator is operational. One hour of error-free operation was veri-
fied in each case. In order to test other compensator CDs, either 5km or 10 km of SSMF
with a −342 ps/nm piece of DCF was taken out from the link. Compensator control was
always successful, and error-free transmission was always possible. Corresponding BER
data (258 km, 253 km) is also plotted in Figure 2.33. The combined penalties of transmis-
sion and CD compensator were measured to be between −1.2dB (an improvement) and
+1.2dB. Figure 2.34 shows received eye diagrams for for NRZ-DPSK and CSRZ-DPSK
modulation formats.
38 Chapter 2 oDPSK Transmission System
2.5 Conclusion
The CD detection scheme is extremely cheap to implement, features superior sensitivity, is
fast enough, introduces hardly any transmission penalty, tolerates NRZ and RZ ASK and
DPSK modulation formats and may be many more modulation formats, provides also the
sign of CD and is believed to have widest measurement range. The 40 Gbit/s CSRZ-DPSK
system also features lock-in stabilized interferometer phase difference and a standard NRZ
clock recovery.
In [72], CD has been compensated at 43 Gbit/s, but for CSRZ-ASK. When that CD
compensator was operational, the Q factors were about 12.5 dB, which was very close to
the FEC limit. Here, the residual chromatic dispersion has been compensated in a 263 km
fiber link at 40 Gbit/s for the modulation formats NRZ-DPSK and CSRZ-DPSK. The syn-
chronous arrival time detection scheme measured residual CD, which was in turn elimi-
nated by automatic control of a −300 ps/nm to −700ps/nm thermally tunable dispersion
compensator. Q factors are >19 dB which corresponds to error free transmission. The
total measured penalty of transmission and CD compensation was −1.2dB ... +1.2dB,
for various link lengths and compensator CDs. CS(RZ)-DSPK outperforms the other mod-
ulation formats in receiver sensitivity, which recommends it for long and ultra-long haul
optical transmission.
Chapter 3
oDQPSK Transmission System
With demands to increase capacity, increase reach and reduce cost, there has been growing
interest in developing alternative modulation formats for high bit rate optical transmission
systems [24–28]. A simple alternative to double the existing transmission capacity or spec-
tral efficiency without optical bandwidth increase is to use differential quadrature phase
shift keying (DQPSK) signals. Combined with RZ coding its robustness against XPM is
also large because the intensity is not modulated by the data but is rather modulated by
pulse carving. The theoretically possible receiver sensitivity for DQPSK signals is better
than for intensity modulated signals. Practically, sensitivity of DQPSK receivers is almost
the same as for the ASK receivers. For the given bit-rate, DQPSK has the symbol rate
which is half of the bit-rate, resulting in increased tolerance to chromatic dispersion and
polarization mode dispersion and reduced spectral and bandwidth requirements for the op-
tical transmitter and receiver. It also known that DQPSK signal tolerates strong optical
filtering [29]. In particular, DQPSK has recently received intense attention for several and
obvious reasons. In this work, DQPSK is explored as an alternative optical modulation
format that has multi-level phase modulation combined with direct detection. Later on,
the DQPSK spectral efficiency was once more doubled using the polarization multiplexing
technique.
3.1 Introduction to oDQPSK
Digital modulation formats are generally characterized by a so called constellation which is
graphical representation of the real and imaginary part of the complex envelope of the mod-
ulated carrier. In optical communication, we have for the representation of the modulated
carrier:
E(t) = Re[a(t)eφ(t)eωc(t)](3.1)
where a(t)eφ(t)is the complex envelope which contains both, amplitude modulation a(t)
and/or phase modulation eφ(t). Optical carrier frequency is determined by the laser center
wavelength ωc. This complex envelope carries the information, allowing phase (or fre-
quency) modulation as well as amplitude modulation.
39
40 Chapter 3 oDQPSK Transmission System
In digital transmission, the bbit are transmitted with a bit rate of 1/Tbbps, where Tb
is the bit duration. In multilevel modulation schemes, the bits are collected and mapped to
digital symbols which are chosen from an alphabet
d(i)∈ {d0, d1, ..., dM−1}, M = 2m(3.2)
of M complex symbols at each symbol interval Ts=mTbnumbered by integer i. Thus, for
4-level PSK (DQPSK) transmission, we have m= 2 and
d0= 1, d1=j, d2=−1, d3=−j. (3.3)
Re{E}
Im{E}
Figure 3.1: DQPSK Constellations
Figure 3.1 shows the DQPSK constellation. In DQPSK modulation format, the infor-
mation is encoded in four different phase states 0, π/2, π, 3π/2 of the optical carrier. The
carrier can assume one of four phases, each change of phase, or symbol, representing 2
bits. The bit combinations being 00, 01, 11 and 10. Consequently, the data stream can
carry 2 bits at a time. Two bits are mapped onto one transmitted optical symbol, offering
bandwidth reduction for increased spectral efficiency. The table 3.1 below illustrates the
mapping from input symbol to output phase transition for DQPSK signal.
Table 3.1: Phase states for DQPSK signal
DataBits PhaseChange
00 0
10 π/2
11 π
01 3π/2
Recently number of experiments are reported using the DQPSK as the modulation for-
mat for high bit rate optical transmission systems having higher spectral efficiencies. Ta-
ble 3.2 summarizes the selected DQPSK transmission experiments having higher spectral
efficiencies and the corresponding transmission distances.
3.2 oDQPSK signal generation 41
Table 3.2: Selected DQPSK transmission experiments with higher spectral efficiencies
Number of Channels Data Rate Distance Fiber Type Efficiency Reference no.
Channels (Gb/s) (km) (b/s/Hz)
40 160 124+200 SSMF+NZ-DSF 1.49 [73]
16 160 153+120 SSMF+NZ-DSF 1.49 [74]
8 160 170+60 SSMF+NZ-DSF 1.6 [75]
64 85.4 320 SSMF 1.6 [76]
50 85.4 300 NZ-DSF 1.14 [73]
8 40 200 SSMF 1.6 [27]
8 20 310 SSMF 0.8 [26]
9 25 1200 SMF-28 0.8 [25]
3.2 oDQPSK signal generation
To perform oDQPSK transmission, one requires a digital precoder, an optical encoder, and
an optical decoder as shown in Figure 3.2.
DFB
τ
τ
ENCODERPRECODER DECODER
di
gi
di
gi
Ii
Qi
Figure 3.2: Schematic representation of Optical DQPSK signalling
3.2.1 DQPSK Precoding
Because of the differential nature of decoding in oDQPSK transmission, a precoding func-
tion is required, as illustrated in Figure 3.2, to provide a direct mapping of the data from
input to output. Mathematically, the operation of the precoder is described by the following
set of equation [24]:
Ii= (Qi−1⊕Ii−1)(di⊕Ii−1)+(Qi−1⊕Ii−1)(gi⊕Ii−1)
Qi= (Qi−1⊕Ii−1)(gi⊕Ii−1)+(Qi−1⊕Ii−1)(di⊕Ii−1)(3.4)
where ⊕denotes exclusive OR, diand giare the original information data bits, and Iiand
Qiare the precoded data bits.
42 Chapter 3 oDQPSK Transmission System
With the precoding function given by (3.4), the output data streams from the decoder
are identical to the data streams input to the precoder: oDQPSK provides optical 2:1 mul-
tiplexing and 1:2 demultiplexing at the optical level. In our lab oDQPSK transmission
experiments as the 27−1PRBS was transmitted, the precoding function was neither im-
plemented not needed.
3.2.2 Optical Encoder
There are several possibilities to generate the optical DQPSK signals using the various
types of devices. Most of them either uses two parallel Mach-Zehnder modulators (MZM)
placed inside the another Mach-Zehnder interferometer forming a Mach-Zehnder super-
structure having quadrature control electrodes to generate four phase states or uses a se-
ries combination of MZM and a phase modulator (PM) to generate four phase states of
the oDQPSK signal. Another possibility is to use only PM driven in such a way that it
again produces the four required phase states. Fourth possibility is to use single dual-drive
modulator to produce nearly arbitrary phase and/or amplitude modulation onto the optical
modulators. Next subsection gives the brief introduction to these methods and as well as
the method that was used in the laboratory to generate the oDQPSK signal.
Parallel Structure using two Mach-Zehnder Modulators
The most widely used method to generate oDQPSK signal is shown in Figure 3.3 [24,62].
The transmitter consists of two parallel Mach-Zehnder modulators (MZM). It requires two
bias controls for each of the MZMs and an active phase control for the phase shifters.
Each of the MZMs of Figure 3.3 is biased for minimum DC transmission and driven with
NRZ data with peak-to-peak amplitude of 2Vπ. Quadrature phase control is required to
recombine signals from the in-phase component (I) and the quadrature component (Q)
with a relative phase difference of π/2. If the two normalized independent drive signals of
such a transmitter are V1and V2, respectively, the baseband complex representation of the
output of the transmitter is the complex number of the form
V1+V2e−j(π/2).(3.5)
The output electric field is thus
E=Re{(V1−jV2)eωct}
=V1cos(ωct) + V2sin(ωct)(3.6)
where Re denotes the real part of a complex number and ωcis the angular frequency of the
optical carrier. Since the output signals of the two Mach-Zehnder modulators are combined
with a π/2phase shift, no coherent superposition occurs in the output Y combiner. The
half of the power is being radiated into the next higher order mode. The same splitting loss
occurs also in the input Y fork [77]. Thus, a 3 dB intrinsic loss exists. The signal exiting
the output of such transmitter is referred to as NRZ-DQPSK signal.
3.2 oDQPSK signal generation 43
electrical
input V2
light in
DQPSK modulated
light out
electrical
input V1
90°
Re{E1}
Im{E1}
Re{E}
Im{E}
Re{E2}
Im{E2}
Figure 3.3: DQPSK signal generation using two Mach-Zehnder modulators
Serial Structure Using Phase and Mach-Zehnder Modulator
A series arrangement of in-phase modulator and a MZM, as shown in Figure 3.4, is also
possible, and has been used in experiments of [26, 28] to generate four phase states of
DQPSK signal. Usually, the first in-phase modulator is driven by NRZ data stream and its
modulation voltage is set to Vπ/2to generate π/2phase shift. The second Mach Zehnder
(MZM) modulator is driven by another NRZ data stream and also performs phase modu-
lation. The modulation voltage of this second modulator is set to 2Vπto generate πphase
shift. There is no intrinsic loss. The relative amplitude error in the in-phase and quadrature
parts of the normalized field in Mach Zehnder modulator can be neglected in comparison
with the phase error in simple phase modulator. The signal that exits the second MZM is
again equivalent to NRZ-DQPSK signal.
DQPSK modulated
light out
electrical input
V1
optical input
electrical input
V2
Re{E1}
Im{E1}
Re{E}
Im{E}
Figure 3.4: DQPSK signal generation using Mach-Zehnder and phase modulator
44 Chapter 3 oDQPSK Transmission System
Single Phase Modulator
A simple phase modulator shown in Figure 2.6 and explained in subsection of optical
phase modulation in chapter 2 can also be used to generate the DQPSK signal. In this
special case, peak-to-peak drive amplitude of one the NRZ data signal is set equal to twice
the peak-to-peak drive amplitude of the another NRZ data signal. The sum of these two
driving signals in fact drive the phase modulator. The phase modulator is biased such that
it produces the optical output signal that takes one out of four phase states of DQPSK
signal φk∈ {0, π/2, π, 3π/2}where input NRZ data streams (k= 1,2) are differentially
precoded. There is no intrinsic loss.
Single dual-drive modulator
A single dual-drive Mach-Zehnder modulator structure may also be used as device for
producing nearly arbitrary phase and/or amplitude modulation onto an optical carrier.
electrical input, V1
light in
modulated light out
electrical input, V2
Figure 3.5: Single dual-drive MZM for DQPSK signal generation
This two arm structure usually results into the output complex envelope:
Eout =Ein
2[exp(π V1
Vπ
) + exp(π V2
Vπ
)] (3.7)
where Vπis the voltage to provide a πrad phase shift of each phase modulator. In the most
trivial case, the MZM is operated as a phase modulator if V1=V2. Thus, by a proper
choice of the both driving voltages V1and V2, any quadrature signal can be generated and
a variety of constellations can be achieved.
The equation (3.7) can be rewritten in the normalized form as
Eout =rmax
2(exp(φ1)−exp(φ2)) (3.8)
where φ1=πV1/Vπand φ2=πV2/Vπ+π. The output electric field, Eout, is the difference
of the two vectors in the circle having a radius of rmax/2. The MZM shown in Figure 3.5
is biased at the point of minimum transmission and the maximum output electric field has
3.2 oDQPSK signal generation 45
an amplitude rmax when V1=V2or φ1and φ2have antipodal phases. The equation (3.8)
gives a geometric representation of the operation of a dual-drive MZM with two indepen-
dent phase modulators. Assume an M-ary signal constellation that can be represented as
complex numbers of the form
si=riejθi, ri>0,0≤θi<2π, i = 1,2, ..., M −1(3.9)
with a maximum amplitude of
rmax =max{r0, r1, ..., rM−1}.(3.10)
With two phases of [78]
φi1=θi+cos−1ri
rmax (3.11)
φi2=θi−cos−1ri
rmax +π, (3.12)
we obtained
si=rmax
2(exp(φi1)−exp(φi2)) .(3.13)
The procedure to find the two phases of φi1and φi2in the circle having radius of
(1/2)rmax for the constellation point of siis described in [78], where the real number
of riis equal to the sum of two conjugated symmetrical complex numbers of (1/2)ri±yi
in the circle with a radius of (1/2)rmax, i.e. (1/4)r2
i+y2
i= (1/4)r2
max. With ϕi=
cos−1(ri/rmax), it is derived (1/2)ri±yi= (1/2)rmax exp±ϕi. Figure 3.6 represents the
two complex numbers of (1/2)rmax exp±ϕias two vectors with phase angles of ±ϕi. The
real number of rigiven by
ri=1
2rmaxejϕi−1
2rmaxej(π−ϕi).(3.14)
All constellations points of (3.9) can be generated based on two phase modulators hav-
ing the phases of (3.11) and (3.12), respectively. The dual-drive MZM in Figure 3.5 can
be used to generate DQPSK signals with constellations of Figure 3.1 when operated as a
phase modulator when φi1and φi2are antipodal phases. The four phases of Figure 3.1
are generated by a four-level drive signal. It is also possible to generate a DQPSK signal
constellation with a smaller number of levels if the four constellation points are reached
with two different two-level drive signals. The scheme that use two-level drive signals has
intrinsic loss of 3 dB [77]. The peak-to-peak drive voltages of the two phase modulators in
Figure 3.5 are proportional to the maximum phase difference of φ1or φ2, respectively. The
maximum phase difference of the four level drive signals is 3π/2and that of the two-level
drive signals is π[78].
46 Chapter 3 oDQPSK Transmission System
π−ϕi
−ϕi
ϕi
φi1=θi+ϕi
φi2
ri
ri
ri e jθi
ϕi
θi
θi
ri e jθi
Radius of
½rmax
Figure 3.6: Procedure to find ϕi1and ϕi2for si=riejθi.
Serial Structure Using MZM and Interferometer
Figure 3.7 shows the laboratory implementation of the DQPSK transmitter using series
combination of single dual-drive MZM followed by the low-cost all-fiber Mach-Zehnder
interferometer (MZI) having differential delay of three symbol durations. This delay is high
enough for decorrelating the data streams but avoids vibration and laser linewidth-induced
differential phase fluctuations.
Re{E}
Im{E}
DQPSK modulated
light out
DPSK modulated
light out
Re{E1}
Im{E1}
Delayed arm
electrical input, V1
light in
electrical input, V2
Figure 3.7: DQPSK signal generation using a dual-drive Mach-Zehnder modulator and interferometer
Mach-Zehnder interferometer with a π/2phase shift in one of the arms converts the
NRZ-DPSK signal generated by the dual-drive modulator to NRZ-DQPSK signal. A piezo
fiber stretcher is included in one of the arms for an active phase control. The measured
polarization-dependent phase shift of MZI is <500 MHz and the extinction ratio is about
24 dB.
3.2 oDQPSK signal generation 47
3.2.3 40 Gbaud DQPSK Transmitter
Figure 3.8 shows the RZ-DQPSK 40 Gbaud transmitter [30,32]. 2×40 Gbit/s DQPSK
signals are generated in a subsequent all-fiber temperature-stabilized Mach-Zehnder inter-
ferometer as described in the previous subsection. At one interferometer output, a 192.5
THz optical bandpass filter (BPF), a photoreceiver with a bandwidth of about ≈12 GHz,
and a subsequent RF diode detector were used to measure the RF power carried by the op-
tical signal. When the two optical signals are superimposed in phase quadrature, there is no
interference and hence no RF power. A quadrature control loop based on a 10 kHz lock-in
detection scheme stabilizes the interferometer phase by minimizing the RF power carried
by the optical signal. The 10 kHz phase modulation has a depth of ∼0.01 rad (rms).
The laser frequencies are fine-tuned to points of a 6.76 GHz ≈1/(2τ)raster so that each
WDM channel contains a proper DQPSK signal. The channel spacing is roughly an odd
multiple of the raster point spacing. This means that each WDM channel has at least one
neighbor whose in-phase and quadrature data streams are combined with opposite polarities
and hence form a different optical pattern. After differential interferometric demodulation
in the receiver, this means that the in-phase and quadrature data streams are exchanged.
Interference
detection
~75 ps
interferometer
10
-
kHz
Lock-in stabilization
Figure 3.8: 2×40 Gbit/s DQPSK Transmitter
In the transmitter, another dual-drive modulator driven at half the clock rate and biased
at the transmission minimum carves 13 ps pulses and thereby generates the return-to-zero
(RZ)-DQPSK signal for transmission. Figure 3.9 shows “eye diagrams” of the intensity of
NRZ-DQPSK and RZ-DQPSK signals at the transmitter.
48 Chapter 3 oDQPSK Transmission System
Figure 3.9: 40 Gbaud intensity eye diagrams of NRZ-DQPSK (left) and CS(RZ)DQPSK signals (right)
3.3 oDQPSK signal detection
Decoding function is normally performed in the optical domain using the Mach-Zehnder
Delay Interferometer (MZDI). Delay interferometer used for DQPSK demodulation is very
similar to the one that was used to demodulate DPSK signals.
3.3.1 DQPSK Decoding
The decoder structure consist of a pair of Mach-Zehnder interferometers, each with an
optical delay τequal to the symbol period Ts= 2Tb. The differential optical phase between
the interferometer arms is set to π/4and −π/4for upper and lower branches, respectively
(Figure 3.10).
Mach-Zehnder
interferometer differential photoreceiver
with lowpass characteristic
electrical
output
Mach-Zehnder
interferometer
electrical
output
optical
input
optical
amplifier
optical bandpass
filter
differential photoreceiver
with lowpass characteristic
π/4
-π/4
+
-
+
-
Figure 3.10: DQPSK Decoder
Balanced optical photoreceivers are employed in each of the interferometer; each of the
photoreceiver pair used in the interferometer has a bandwidth >50 GHz. If the input signal
has the form Eoe−(ω0t+4φi), then the output signals after balanced detection are propor-
tional to: (cos 4φi+ sin 4φi)and (cos 4φi−sin 4φi), respectively. The output signals
are, therefore, binary NRZ signals. Standard clock and data recovery circuits can there-
fore be used. Generally, an adaptive polarization control is not needed for demodulation
of oDQPSK signals if the receiver interferometer is free from the polarization–dependent
loss.
3.4 2×40 Gbit/s DQPSK Transmission Experiment 49
3.3.2 40 Gbaud DQPSK Receiver
The receiver employs optical preamplifiers, a flat-top C band DWDM DEMUX (Optun)
and an integrated-optical Mach-Zehnder demodulator with a delay of 4 symbol durations
(Figure 3.11). For proper reception of in-phase and quadrature data channels, the phase
difference of delay demodulator is set to π/4or -π/4using differential micro–heaters with
total constant power. The demodulator outputs are connected to two high-speed photode-
tectors (u2t), which are connected to differential inputs of a 1:16 Infineon demultiplexer
that uses standard clock and data recovery circuits. Main advantage of this scheme is that
here we do not need an extra high-speed photodiode to recover the clock from 40 GHz
intensity modulation. A 27−1PRBS data generated using the polynomial 1 + X6+X7
was transmitted. Precoding function described by equation (3.4) was neither needed nor
implemented as the PRBS pattern was transmitted. Therefore, there was a deterministic
mapping of the data from the input to the output. As a result, the demodulated bit patterns
in in-phase and quadrature data channels differ from the transmitted ones. To enable bit-
error-rate (BER) measurements, the error detector was properly programmed to receive the
expected data sequence using the DQPSK mapping.
3.4 2×40 Gbit/s DQPSK Transmission Experiment
The aim of this transmission experiment is to demonstrate 2×40 Gbit/s RZ-DQPSK trans-
mission and compare its performance to that of the RZ-ASK and RZ-DPSK modulation
formats in terms of receiver sensitivity and OSNR.
3.4.1 Transmission setup
Figure 3.10 shows the RZ-DQPSK 40 Gbaud transmission setup. For RZ-DPSK operation,
the all-fiber Mach-Zehnder interferometer at the TX is left out. For RZ-ASK operation,
both interferometers and one photodiode are left out.
The optical signal is transmitted over 3fiber spans with a total length of 263 km. These
three spans consist of 170 km of standard single mode fber (SSMF), 60 km of nonzero
dispersion shifted fiber (NZDSF), and 33 km of dispersion shifted fiber(DSF). Dispersion
compensating fiber (DCF) with a total dispersion of −2713 ps/nm was inserted between
first and second stages of the two inline EDFAs. A thermally tunable dispersion compen-
sator from TeraXion compensates for the residual dispersion of the 192.5THz channel.
Dispersion was set to −470 ps/nm while the total tuning range of the dispersion compen-
sator is from −300 to −700 ps/nm.
50 Chapter 3 oDQPSK Transmission System
100 ps
20 GHz
RZ
75
ps
10 kHz
lock-in
10 GHz
PIN-TIA
DQPSK
DCFDCF
DFB
laser DPSK
5 MHz
FM&AM 40 Gbit/s
data
out
clock
and data
recovery
loop
filter
clock
VCO
tunable CD
compensator
90
km
84
km
89
km
RF power
detector
Figure 3.11: 2×40 Gbit/s RZ-DQPSK transmission setup
3.4.2 Measurement Results and Discussion
Figure 3.12 shows measured bit-error-ratios (BERs) vs. optical preamplifier input power in
dBm for RZ-DQPSK, RZ-ASK, and RZ-DPSK modulation formats. The back-to-back
Q factors for these modulation formats are 20.9dB (for both I and Q data channels),
26.6dB, and 29.5dB, respectively. The corresponding back-to-back receiver sensitivi-
ties are −27.5dBm (for both I and Q data channels), −27.3dBm, and −33.6dBm. They
are equivalent to OSNRs of 29.7,27.7, and 23.8dB/0.1nm, respectively. With the 263 km
fiber link in place, the Q factors are reduced to 17.5dB (for I and Q data channels), 19.6,
and 20.4dB, respectively.
As can be seen from Figure 3.12, the DQPSK receiver sensitivity is almost the same as
for ASK. However, DQPSK transports 80 Gbit/s whereas ASK transports only 40 Gbit/s.
In principle, the similar bit rates can be also achieved, for example, by using the multilevel
modulation formats based on amplitude shift keying (ASK) [79,80]. This class of signals
is known as M-ary ASK signals.
The transmission bandwidth of a multilevel ASK signal (M-ary signal), where Mis
the number of levels in the M-ary signal, is scaled by a factor 1/log2(M)compared to a
binary signal operating at the same bit rate. Similarly, the symbol period is increased by
a factor log2(M)compared to a binary signal. The back-to-back sensitivity, however, is
significantly degraded for these type of signals because of the increased number of levels
in signals and the signal dependence of signal-spontaneous beat noise [81].
3.4 2×40 Gbit/s DQPSK Transmission Experiment 51
-40 -38 -36 -34 -32 -30 -28 -26
10-11
10-9
10-7
10-5
10-3
RZ-DPSK
RZ-ASK
BER
I & Q
RZ-DQPSK
Sensitivity [dBm]
Sensitivity [dBm]
Figure 3.12: Measured BERs vs. optical preamplifier input power for RZ-DPSK, RZ-DQPSK, RZ-ASK
In a presence of stationary noise only, and assuming a white noise spectral density, the
additional optical power required for detecting an M-ary signal over a binary signal signal
is [81]
PM=M−1
plog2(M).(3.15)
According to equation (3.15), a 4-ary ASK requires about 3.3dB more optical power than
a conventional binary signal.
The measured receiver sensitivity (Figure 3.12) is still better for RZ-DPSK signals, but
the main advantage is that RZ-DQPSK signal simply doubles the transmission capacity.
When the sensitivities are compared on the basis of photons/bit (not photons/symbol) then
DQPSK is 3.2dB better than ASK, and 3.1dB worse than DPSK. All 2.5Gbit/s sub-
channels are bit error free, with the almost identical sensitivities.
Figure 3.13 shows 2×40 Gbit/s RZ-DQPSK eye diagrams back-to-back (top) and af-
ter 263 km transmission (middle) for in-phase and quadrature reception. The bottom eye
diagram with 3lines results when the interferometer phase difference was set to either 0◦
or 90◦instead of 45◦or 135◦for DQPSK reception. The eye diagrams are well open, both
back-to-back and after transmission over 263 km of fiber. The in-phase and quadrature
data channels were tested (in 1out of 16 2.5Gbit/s sub-channel) to be error-free during
1h each, but these measurements were interrupted (before errors occurred) for occasional
phase adjustment in the receiver interferometer, and polarization adjustment. Shorter trans-
mission spans were also tried, 258 km (5km less SSMF) and 253 km (10 km less SSMF
but increased link dispersion because a −342 ps/nm DCF module was also taken out).
52 Chapter 3 oDQPSK Transmission System
Figure 3.13: 2×40 Gbit/s RZ-DQPSK I and Q eye diagrams back-to-back (top) and after 263 km of fiber
(middle). Bottom diagram is back-to-back with wrong interferometer phase
Error-free transmission was possible, though not extensively tested. The chromatic
dispersion compensator had to be set to −390 ps/nm (for 258 km) and −635 ps/nm (for
253 km), respectively. In those cases as well as for 263 km (see Figure 3.13) the eye
diagrams before and after transmission had identical shapes, which suggests that the com-
pensator did not introduce a significant penalty. This is remarkable because DQPSK is
more sensitive to chromatic dispersion than DPSK or ASK. However, as the signal is trans-
mitted, the in-phase part of optical amplifier noise modulates the pulse amplitudes. Self
phase modulation converts this into a random phase modulation which limits permissible
link lengths. This nonlinear phase noise is also discussed in [82,83]. Although, this is not
strictly necessary because linear phase noise is generally included as a part of any sensi-
tivity calculation in which optical amplifier noise is taken into account. It scales with the
square of the length and linearly with the symbol rate (taking into account that the linewidth
tolerance scales also linearly with the symbol rate). Launch powers were 4...6dBm for the
3 spans in the present experiment, and the laser linewidth is <2MHz according to the
Triquint data sheet.
The setup could be made less sensitive against phase noise if the interferometer delay
in the receiver were shortened to 1 bit. In theory, this should at least double the permissible
transmission distance but we don’t know precise experimental limits yet. Definitely, the
use of Forward Error Correction (FEC) technology, in practice, will relax the problem.
3.5 RZ-DQPSK Polarization Multiplex Transmission 53
3.5 RZ-DQPSK Polarization Multiplex Transmission
DQPSK [24–26,28,31,32] and polarization division multiplex (PolDM) [33] transmission
each can double the fiber capacity by their increased spectral efficiency. Both techniques
have been combined to transmit 4×10 Gbit/s per WDM channel [27,62].
In this work, a 160 Gbit/s (4×40 Gbit/s) transmission system is realized by combining
DQPSK with polarization division multiplex, for the first time at a line rate of 40 Gbaud.
The fiber capacity equals 1.6bit/s/Hz, which value has previously been achieved or sur-
passed only at 10 Gbaud [27,62].
3.5.1 Transmission Setup
Figure 3.14 shows the RZ-DQPSK polarization division multiplex (PolDM) 4×40 Gbit/s
per WDM channel transmission setup [75]. Eight 100-GHz spaced WDM signals (192.3 ...
193.0 THz) are combined with equal polarizations and are modulated together.
20
GHz
RZ
10 kHz
lock-in
loop
filter
clock
VCO
data out
clock
and data
recovery
25 ps
tunable CD
compensator
RF power
detector
BPF
DCF
230 km of
SSMF and NZDSF
in 4 spans
75
ps
DQPSK
DPSK
192.3 THz
193.0THz
LiNbO3polarizer
controller RF power
detector
PolDM
40 Gbit/s
27-1
Figure 3.14: 4×40 Gbit/s per channel RZ-DQPSK PolDM transmission
First, 2×40 Gbit/s DQPSK signals are generated (Section 3.4). In order to increase
the bit rate from 80 to 160 Gbit/s per WDM channel, an existing polarization division mul-
tiplexer (PolDM) was employed. PolDM is a quaternary modulation scheme where one
bit modulates the horizontal and the other bit modulates the vertical electric field [33]. It
doubles the data rate in existing trunk lines without need for an additional optical band-
width. Thus, the DQPSK signal is split and recombined with orthogonal polarizations with
a differential delay of 2.8ns. Since this polarization multiplexer (PolDM) was available,
interleaving of orthogonally polarized pulses in the time domain was not tested.
54 Chapter 3 oDQPSK Transmission System
The optical signals are transmitted over ∼230 km of fiber in 4fiber spans having
∼170 km of SSMF and ∼60 km of NZDSF. DCF with dispersions of −1345,−685, and
−683 ps/nm is inserted in between two inline EDFAs, in pairs, respectively (−2713 ps/nm
in total). Fiber and DCF launch powers are −0.5. . . + 4 dBm and −4.8. . . −3dBm
per WDM channel, respectively. EDFA input powers are −15 . . . −10.5dBm per WDM
channel.
The receiver contains optical preamplifiers and a flat top C band DWDM DEMUX. To
receive the 192.5THz (1557.366 nm) channel, the TeraXion thermally tunable dispersion
compensator is set to −440 ps/nm. Group delay vs. wavelength for various dispersion set-
tings has been already shown in Section 2.4 (Figure 2.31). Other WDM channels are not
compensated because only a single channel TDC [69] was available. Automatic polariza-
tion control is implemented in the receiver to recover both polarizations. A LiNbO3–based
polarization controller is followed by a polarizer. The control strategy is again based on the
minimization of the broadband RF interference noise. It occurs when both polarizations
are present after the polarizer. A linear ideal polarizer is an optical device, birefringent
or not, that only transmits one linear state of polarization and suppresses any transmission
of the orthogonal state of polarization. Although a real component always lets through a
fraction of the orthogonal state. These device is characterized by a Jones matrix JPwhich
is expressed with respect to a reference coordinate system Oxy. If the phase factor, which
simply renders the propagation of the light in the material medium making up the device,
is not taken into account, the Jones matrices JPxand JPyof the polarizers whose principal
axes are respectively the axes Ox and Oy are given by:
JPx=1 0
0 0 JPy=0 0
0 1 (3.16)
Polarizer is used to suppress the orthogonal polarization and to ensure that phase modulated
light having a single polarization enters the Mach-Zehnder delay interferometer for demod-
ulation. The interference noise is detected in another 12 GHz photoreceiver followed by
an RF power detector (Figure 3.15). The measured RF power is −22 dBm in the best case
(when the two polarizations are well aligned) and −8.5dBm in the worst case (when both
polarizations pass the polarizer with equal powers). The controller tries to minimize the
interference noise by suppressing the unwanted polarization in the fiber polarizer. Signal
acquisition takes around 1 s, and this is fast enough to track occurring fiber polarization
changes.
Another Mach-Zehnder interferometer, with a delay of one symbol duration, demodu-
lates the signal. For proper reception of in-phase and quadrature data channels, the phase
difference of the delay demodulator is set either to 45◦or 135◦, using a piezo fiber stretcher.
The demodulator outputs are connected to two high-speed photodetectors, which in turn are
connected to the differential inputs of a 1:16 demultiplexer with standard clock and data
recovery. Note that the demodulated bit patterns in in-phase and quadrature data channels
differ from the transmitted ones. The half rate clock signals in transmitter and receiver are
generated by VCOs from WORK Microwave GmbH.
3.5 RZ-DQPSK Polarization Multiplex Transmission 55
0 2 4 6 8 10 12 14 16
-80
-60
-40
-20
0
Frequency [GHz]
Spectral power [dBm]
-120
-100
-80
-60
-40
Figure 3.15: Electrical interference spectra measured in the 12 GHz photoreceiver after the polarizer
3.5.2 Transmission Results
Figure 3.16 shows the recorded back-to-back sensitivities of the 4×40 Gbit/s, 192.5THz
signal, for which the TDC was operational. For a BER of 10−9the sensitivity is about
−22 dBm. At the forward error correction (FEC) threshold, say for a BER of 10−3, it is
about −32 dBm.
BER
Sensitivity [
dBm
]
Y-Pol. (I)
X-Pol. (I)
Y-Pol. (Q)
X-Pol. (Q)
-
32
-
30
-
28
-
26
-
24
-
22
-
20
-
18
10 -11
10 -9
10 -7
10 -5
10 -3
Sensitivity [dBm]
Figure 3.16: Back-to-back receiver sensitivity for both in-phase and quadrature data channels for one polar-
ization. Optical power is given for aggregate 160 Gbit/s signal
Figure 3.17 shows measured back-to-back Q factors, calculated from BER measure-
ments, for I and Q data streams for all 8 WDM channels. A Q ≥15.6dB or BER ≤10−9
is achieved for all channels, polarizations and quadratures. After transmission over 230 km
of fiber, a BER ≤10−9is obtained for the 192.5THz channel with CD compensation.
Corresponding data, expressed as Q factors, is also given in Figure 3.17.
56 Chapter 3 oDQPSK Transmission System
Y-Pol. (Q)
Y-Pol. (I)
X-Pol. (I)
X-Pol. (Q)
24 26 28 30 32 34 36
11
12
13
14
15
16
17
Q [dB]
OSNR [dB/0.1nm]
Figure 3.17: Back-to-back performance of 4×40 Gbit/s system
The eye diagrams corresponding to back-to-back configuration and after transmission
over 230 km of fiber are shown in Figure 3.18. The case of other polarization is very simi-
lar. The eye diagrams before and after transmission have identical shapes, which indicates
a clean transmission with effective CD compensation. This is remarkable because an ex-
trapolation of the results in [27], and our own experience, tells that DQPSK tolerates less
chromatic dispersion than DPSK at the same symbol rate.
Figure 3.18: Eye diagrams in one polarization, (top) back-to-back in I channel, Q channel and (bottom )after
230 km in I and Q channel
3.5 RZ-DQPSK Polarization Multiplex Transmission 57
In Figure 3.19 the optical spectrum after 229 km of fiber is shown.Figure 3.20 shows
Q factors, directly calculated from the measured BER values, for the back-to-back case
against the OSNR. The OSNR is determined in an 0.1nm bandwidth by comparing the
spectral peak against the surrounding noise. To reach Q = 15.6dB the required OSNR is
about 33 dB. At the FEC threshold the required OSNR is about 22 dB.
1550 1552 1554 1556 1558 1560
-70
-60
-50
-40
-30
-20
-10
0
Power[dBm]
Wavelength [nm]
Figure 3.19: Optical spectrum after 229 km of fiber
The presence of the other WDM channels confirms that the capacity is 1.6bit/s/Hz. Si-
multaneous BER measurement of all WDM channels would require a broadbanddispersion
compensator [25]. If FEC is available, amplifier spacing and/or WDM channel number are
expected to be expandable.
10
15
20
15
20
25
30
10
192.2 192.4 192.6 192.8 193.0
Q [dB] in X pol.
Frequency [THz]
, = back-to-back I, Q
192.2 192.4 192.6 192.8 193.0
, = 230 km I, Q @ 192.5 THz
25
5
Q [dB] in Y pol.
Figure 3.20: Measured Q factors for I and Q data channels in both polarizations back-to-back for 8WDM
channels, and after transmission over 230 km fiber for the CD-compensated 192.5THz channel
System stability was limited to ∼1min due to insufficient thermal isolation of the
receiver interferometer. Recently, the receiver interferometer has been packaged with the
58 Chapter 3 oDQPSK Transmission System
commercial styrofoam, which has drastically improved the system stability. However, long
term stability has not yet been assessed.
By using only 100 GHz channel spacing, a 1.6 bit/s/Hz spectral efficiency is achieved.
In [28], a 70 GHz spacing was used for 2×42.7Gbit/s DQPSK transmission. Combining
such a channel spacing with polarization division multiplex should make spectral efficien-
cies beyond 2bit/s/Hz possible.
3.6 Conclusion
The 2×40 Gbit/s RZ-DQPSK error-free signals are transmitted over a 263 km fiber link.
A40 Gbit/s tunable chromatic dispersion compensator and a standard 40 Gbit/s DWDM
DEMUX are used; fiber capacity is simply doubled. The receiver sensitivity is -27.5 dBm.
The back-to-back Q factor is >20 dB. Even after transmission the Q factor is 17.5dB.
2×40 Gbit/s RZ-DQPSK transmission over a 263 km fiber link was reported. Sufficient
resilience against nonlinear phase noise and band limitation in a 40Gbit/s WDM DEMUX
is achieved by a Q factor of 17.5dB. The receiver sensitivity of −27.5dBm is 0.2dB better
than for RZ-ASK and 6.1dB worse than for RZ-DPSK but the data rate is twice as high.
For the first time, a 160 Gbit/s (4×40 Gbit/s) DQPSK on each of 8 100 GHz-spaced
WDM channels using a 40 Gbit/s tunable chromatic dispersion compensator and a standard
40 Gbit/s DWDM DEMUX has been demonstrated. Data is carried in two polarizations and
differentially encoded in two quadratures. Fiber capacity per WDM channel is therefore
quadrupled. A 1.6bit/s/Hz transmission over 230 km of fiber is achieved with Q >15.6dB
for one of the 8WDM channels for which the tunable dispersion compensator was opera-
tional.
Chapter 4
High-Speed Integrated Circuits for
oDPSK Transmission
High-performance low-cost physical layer integrated circuits are needed for the successful
implementation of next generation 40 Gbit/s optical networks. To expand the transmis-
sion capacity of the existing wavelength division multiplexed (WDM) networks, such high
bit rates must have to be realized at the single channel level. This prerequisite imposes
significant technological demands on the optical front ends. To date, the high-speed (i.e.
10 and 40 Gbit/s) demonstrations in the literature mainly focuses on the transmitter hard-
ware [36–40,84,85] as opposed to receiver hardware. In this work, a differential amplifier
combined with Travelling Wave Amplifier concept is simulated in GaAs technology while
differential in and differential out linear amplifier is demonstrated in 0.18 µm CMOS tech-
nology using striplines.
4.1 Differential Amplifier for 10 and 40 Gbit/s CS(RZ)-
DPSK system
To build the commercial (CS)RZ-DPSK receiver, the architecture of the conventional NRZ
optical receiver must be changed. It has been shown both theoretically and experimentally
that roughly a 3 dB improvement in system margin can be achieved by using a balanced
optical front end (OFE) instead of a single-ended OFE (see Figure 4.1). Therefore, for high-
speed systems using CS(RZ)-DPSK or DQPSK, particularly at 10 and 40 Gbit/s, the design
of an integrated balanced optical front end (OFE) can be extremely challenging due to
packaging issues and integrated circuit performance which basically includes differential in
and differential out linear amplifier with two matched photodiodes, and is thus quite worthy
of significant attention. Such optical front ends are needed to reach record sensitivity limit
of about −35.5dBm or less and optical signal-to-noise ratio (OSNR) performance of or
around 18.5 dB in a 0.1 nm bandwidth at a BER of 10−9. Such a sensitivity or OSNR
performance is very difficult to achieve without using the linear differential amplifier.
59
60 Chapter 4 High-Speed Integrated Circuits for oDPSK Transmission
Differential processing of input RF signals removes some waveshape distortions result-
ing from RF group delay variations and amplitude distortions. Typically, the common mode
rejection ratio (CMRR) should be better than 20 dB [86]. The methodology that is outlined
here could provide a commercial path without the use of monolithic integrated circuits.
Such an approach usually takes an advantage of the hybrid technologies that allows us to
use better photodiodes and differential amplifier performance because of the individually
optimized fabrication processes.
100 ps
MZI
DPSK
Optical Input
Optical
Preamplifier Differential In /
Differential Out
Linear Amplifier
Hybrid Balanced OFE
oBPF
Figure 4.1: Typical 40 Gbit/s CS(RZ)-DPSK balanced optical front end
Figure 4.1 shows the typical schematic of the CS(RZ)-DPSK balanced optical front
end. Received input optical signal (DPSK) is first passed through the optical preamplifier
followed by the optical bandpass filter (oBPF) to improve the OSNR and then through the
delay demodulator (Mach-Zehnder interferometer) to generate two complementary ASK
signals for direct detection. It is desirable to differentially amplify these directly detected
signals before they are passed on to the standard clock and data recovery circuits in order
to achieve better signal-to-noise ratio performance. The linear differential amplifier could
be used for this purpose.
Typically, such differential in and differential out linear amplifier should have a small
signal gain of around 20 dB, a 3 dB bandwidth of at least 36 GHz, and a nominal maximum
output swing of 400 mV per channel for 40 Gbit/s application [86]. The circuit should be
DC coupled at the input and output and therefore, must have separate input offset voltage
terminals to set the desired output DC offsets for the data and data complement outputs.
Such circuits at 40 Gbit/s could be realized either in GaAs or InP or in SiGe technol-
ogy [86]. CMOS technology could become an alternative for 10 Gbit/s applications and
beyond [45,87–89].
4.2 Differential Distributed Amplifier
Thedifferentialdistributedamplifier presented hereisbasedon the OMMIC D01PH pseudo-
morphic AlGaAs/InGaAs HEMT technology which was specifically developed for power
4.2 Differential Distributed Amplifier 61
applications and operational frequencies up to the millimeter wave region [90]. Typically,
D01PH process exhibits the cut-off frequency in the range of ∼100 GHz.
Vds [V]
Ids [mA]
Figure 4.2: Simulated DC characteristics of the HEMT fabricated in OMMIC D01PH process
Figure 4.2 shows the simulated HEMT (size: 2×40 µm) characteristics of drain-to-
source current, IDS, versus drain-to-source voltage, VDS, for different values of gate-to-
source voltage, VGS. It is possible to operate the HEMT with zero gate-to-source voltage.
The HEMT‘s in distributed amplifying stages are operated with zero gate-to-source voltage
while in the differential amplifying stages they are biased.
4.2.1 Distributed Amplification
The concept of distributed amplifiers dates back to the 1940s when it was used for the
first time in the design of broadband vacuum tube amplifiers. With recent advances in
microwave integrated circuit and device processing technology, the distributed amplifiers
found new applications in broadband microwave amplifiers [91]. Bandwidth in excess of
decade are possible with good input and output impedance matching. But, they are gen-
erally larger in size than amplifiers having a comparable gain over a narrower bandwidth.
Distributed amplifiers are also known as travelling wave amplifiers (TWA).
Most distributed amplifiers since the early 1980’s have been realized as MMIC’s on
compound semiconductor technology (GaAs or InP) [38–42]. Recently, interest in MOS-
FET distributed amplifiers [92–96], has been fueled by the fact that a standard submicron
CMOS process can reach operating speeds well into the microwave range [97]. However,
there is still a considerable obstacle in the realization of useful CMOS distributed ampli-
fiers due to the difficulty in realizing high-quality factor inductors and transmission lines in
a standard CMOS process [93].
The basic configuration of a distributed amplifier is shown in Figure 4.3. The ampli-
fier consist of two transmission lines on the input and the output, and multiple transistors
providing gain through multiple signal paths. The forward (from left to right in Figure 4.3)
wave on the input line is amplified by each transistor. The incident wave on the output line
travels forward in synchronity with the travelling wave on the input line.
62 Chapter 4 High-Speed Integrated Circuits for oDPSK Transmission
RG
RD
In
ld
lg
Transmission
line sections
Out
1••• N
Figure 4.3: Basic configuration of the travelling wave amplifier
Each transistor adds power in phase to the signal at each tap point on the output line and
therefore, the whole amplifier is capable of providing a higher gain-bandwidth product than
a conventional amplifier. The forward travelling wave on the gate line and backward (trav-
elling from right to the left) wave on the drain line are absorbed by terminations matched to
the loaded characteristic impedance of the input line, RG, and output line, RD, respectively,
to avoid reflections. The extended bandwidth of the distributed amplifier comes at the price
of a larger time delay between its input and output, as there is a trade-off between the band-
width and delay in an amplifier. Alternatively, one can think of this approach as a method of
absorbing the parasitic capacitances of the transistors into transmission line structures and
making them a part of the passive network [91,98]. According to the inductive–capacitive
(LC) model of the transmission line as derived in Appendix B, this added capacitance por-
tions reduce the impedance of the gate and drain line to
Z0G≈sL0
g
C0
g+Cgs/lg
Z0D≈sL0
d
C0
d+Co/ld
.(4.1)
As has been mentioned above, the resulting waves on the drain line travel in forward direc-
tion in synchronity with the travelling wave on the gate line. Matched terminations absorb
the forward wave on the gate line and backward wave on the drain line. Thus, the phase
velocities
υph =1
√L0C0(4.2)
on the gate and drain line must coincide. As the input capacitance is normally larger than
the output capacitance, a constructive superposition, for example, can be achieved by using
the different transmission line lengths for the gate and drain line. Another possibility is to
insert the appropriate capacitance either in parallel with the drain line or in series with the
gate line.
4.2 Differential Distributed Amplifier 63
The total TWA gain resulting from Ntransistors stages as has been derived in Appendix
B is given by the following expression:
Gp=g2
koZ0GZ0D[exp(−αGNlg)−exp(−αDNld)]2
4(αDld−αGlg)2.(4.3)
Reducing the number of stages will obviously reduce the gain [99]. The simple solution is
to increase the size of the individual transistor in order to effectively increase the transcon-
ductance, gm, of the individual transistor. However, an increase in the size of transistor also
increases the gate capacitance and hence decreases the cut-off frequency
fc=1
πZ0GCgs
.(4.4)
The number of stages which maximizes the gain of a traveling wave amplifier at a given
frequency can be approximated by [99]
Nopt =ln(αGlg/αDld)
αGlg−αDld
.(4.5)
As a result, the gain of a distributed amplifier cannot become infinity. The overall gain
of the travelling wave amplifier increases with corresponding increase in the distributed
stages until the optimum number of stages has been reached at the given frequency. Any
further increase in the distributed stages beyond the optimum number is not useful because
the signal can not overcome the attenuation in the extra sections of the drain line. As a
result, the gain of the travelling wave amplifier begins to decrease with further increase in
the number of distributed stages.
Cascode Stage
Traveling wave amplifiers normally use the cascode as the main amplifying stage. Fig-
ure4.4is an exampleof a cascode amplifier, a common-sourcetransistor driving a common-
gatetransistor. Acascodeamplifierhas the same overallvoltagegain as that of the common-
source amplifier. The main advantage of a cascode amplifier is its lower input Miller capac-
itance, which is considerably less than the input capacitance of a common-source amplifier.
Compound device also provides a higher output impedance and reduced “reverse internal
feedback”. But the main drawback of the cascode stage is that it exacerbates the stability
problems cased by resonance in the S22 parameter of the amplifier.
4.2.2 Circuit Design
ThedifferentialdistributedAmplifier(DDA) basically consists ofa differentialpreamplifier
circuit with two differential outputs, each driving the gate line of a TWA with four stages
[35].
64 Chapter 4 High-Speed Integrated Circuits for oDPSK Transmission
+Vdd
Vin
Vout
Vb
Rd
Figure 4.4: Typical schematic of the cascode amplifier
Differential Pre-amplifier Circuit
Figure 4.5 (left) shows the schematic of the lumped differential preamplifier which acts as
an input stage of the differential distributed amplifier. Differential pre-amplifiers features
on-chip 50 Ωresistors to provide good impedance matching. The differential input signals
are coupled into the differential preamplifier using the 50 Ωmatched coplanar waveguides.
The width of the central conductor and the gap between the central conductor and the
groundplanes of the coplanar waveguidesare 13 µm and 18.5µm, respectively. Differential
preamplifier has simulated power gain of ≈3dB at 40 GHz while the simulated Common
Mode Rejection Ratio (CMRR) is better than 16.5 dB up to 40 GHz as shown in Figure 4.5
(right).
CMRR [dB]
Freqency [GHz]
0 10 20 30 40 50
40
30
20
10
0
S21[dB]
0 10 20 30 40 50
8
6
4
2
0
Frequency [GHz]
Vss
RL
RL
vOUT
I0
vIN1 vIN2
Vdd
Figure 4.5: Schematic of the differential pre-amplifier (left) and simulated magnitude of S21 and CMRR
(right)
4.2 Differential Distributed Amplifier 65
Travelling Wave Amplifier
Figure 4.6 shows the schematic of the traveling wave amplifier. The circuit is designed
to work with the drain bias voltage of 2.8 V which could be applied through an external
bias tee. This voltage is also used on chip for the drain line termination resistor, avoiding
the DC losses that would result from termination to ground. To obtain larger bandwidth,
a common-source common-gate (cascode) amplifier is used as the main amplifying stage.
Thus, a single cascode stage consists of a pair of two finger HEMTs in a common-source
common-gate cascode configuration where every finger has a gate width of 40 µm. Cascode
is designed as a single cell in order to save space and reduce parasitics. The gate bias
voltage of common-gate HEMT is set to 1.4 V.
In
VdOut
Cgate Rgate
Vb
Single stage ...
Figure 4.6: Schematic of a traveling wave amplifier using cascode as the main amplifying stage
The main drawback of the cascode cell is that it causes the resonance in S22 of the
amplifier, which may lead to stability problems. Therefore, in order to improve stability,
a series-damping resistor at the gate of the common-gate transistor is inserted. Simulation
results confirm that the amplifier is unconditionally stable which is indicated by the both
stability factors (kand µ)>1. Figure 4.7 shows both the stability factors (left) and the
output reflection coefficient, S22, (right) which is below -15 dB up to 40 GHz.
10 Stability Factors, 100
-10
-20
-30
-40
Frequency [GHz]
S22 [dB]
0 10 20 30 400 10 20 30
Frequency [GHz]
k
µ
Figure 4.7: Stability factors kand µ(left) and output reflection coefficient S22 (right)
66 Chapter 4 High-Speed Integrated Circuits for oDPSK Transmission
The characteristic impedance of the coplanar waveguides used in the schematic of trav-
eling wave amplifiers is 65 Ω. The central conductor width and gap between the central
conductor and the ground planes is 8µm and 21 µm, respectively. Top metal is used as a
conductor metal.
Simulated drain and gate line lengths for optimum gain with respect to the desired
bandwidth are 200 µm and 150 µm, respectively. Such longer lengths are permissible using
the meander lines in the layout option, however they complicate the layout and simply
consume more area. As an alternative to longer drain line, a small capacitor of the order
of 0.03 pF was used to slow down the signal. Thus, the gate and drain phase are matched
as evident from the simulation shown in Figure 4.8. Figure 4.9 shows how the capacitor is
integrated in the T-junction of the drain line.
Frequency [GHz]
0 10 20 30 40
200
100
0
-100
-200
Gate and Drain
Phase [°]
Figure 4.8: Phases on the gate and drain line
All resistors in the schematic which are used in simulation use a thin film N+ active
layer with a typical sheet resistance of 100 Ω/square. DC voltages are decoupled on the
chip using the decoupling capacitors having a 150 nm SiN dielectric layer.
gate line
drain line
integrated
capacitor
Figure 4.9: Layout details of the cascode cell
4.2 Differential Distributed Amplifier 67
Frequency [GHz]
0 10 20 30 40
50
40
30
20
10
Group Delay [ps]
Frequency [Hz]
S21 [dB]
N = 4
N = 5
N = 3
Figure 4.10: Optimization of forward transmission as a function of number of stages N(left) and group
delay (right)
Figure 4.10 (left) shows the optimization of forward transmission (S21) versus the stage
numbers (for N= 3 ···5). For N= 4, a forward transmission gain of 17.2dB with a
46.8GHz bandwidth is simulated. The gain at 40 GHz is ∼15 dB. Characteristic response
(gain versus frequency) is very flat. The simulated group delay has a very slight rising
trend with respect to the increase in frequency. The simulated group delay difference in the
frequency range up to 40 GHz is less than 10 ps as depicted in Figure 4.10 (right).
Time [ps]
Output voltage [VPP]
Figure 4.11: Simulated eye diagram for 50 mVpp input voltage
Figure 4.11 shows the simulated eye diagram having the output voltage of around
0.37 Vpp for the input voltage of 50 mVpp. The simulated gain of the differential amplifier
for 40 Gbit/s PRBS signals is higher than 17 dB.
Figure 4.12: Layout of the differential distributed amplifier at 40 Gbit/s
68 Chapter 4 High-Speed Integrated Circuits for oDPSK Transmission
Figure 4.12 shows the layout of the differential distributed amplifier which has the
dimensions of 3 mm×1 mm. It is realized to fulfill important high-frequency circuit design
goal- a total layout symmetry. The input and output data lines are in GSGSG configuration
with a 100 µm pitch.
4.2.3 Result Disscusion
This section presents the state of the art technique to implement the high-gain differential
amplifiers at 40 Gbit/s. Simulation results indicate that the differential distributed amplifier
has a maximal simulated gain of 17.2 dB in a 3 dB frequency bandwidth of 46.8GHz and
a CMRR of ∼16.5dB at 40 GHz.
Results are comparable with the optical front-end circuit used by Sinsky et al [86].
Their linear differential amplifier was built using a high-performance SiGe process and
uses a traveling wave design. They have achieved 21 dB of small signal gain in a 3 dB
frequency bandwidth of 36 GHz with the CMRR of 20 dB at 40 GHz. Using this circuit
they reported the record sensitivity of -35.9 dBm (39 photons/bit) for a BER of 10−9and
an OSNR of 17 dB in 0.1 nm bandwidth for the reception of RZ-DPSK signals [86].
4.3 10 Gbit/s CMOS Differential Amplifier
Traditionally, high-speed circuits are realized either in GaAs or in InP technology. Re-
cently, the SiGe technology became an alternative to both. However, the above mentioned
technologies have relatively high cost of integration. An approach that will drastically re-
duce the cost is the standard CMOS technology. Another advantage of using the CMOS
technology is that it has a high packaging density and relatively low power dissipation. On
the other hand, it is relatively slow. It is indeed difficult to implement passive structures
because of high propagation loss due to the low bulk-resistivity of CMOS substrates [100].
In this work, a single-stage differential pre-amplifier, followed by three pairs of distrib-
uted common source stages using striplines, is implemented in standard 0.18 µm CMOS
technology. Although, striplines (SL’s) are lossy as compared to microstrips (MS’s) or
coplanar waveguides (CPW’s), SL has a certain advantage over both of them when used in
complex analogue circuits for system integration duo to the perfect shielding. Additionally,
the striplines are completely surrounded by the dielectric material. As a result, they are free
from dispersive effects.
Twomost popular technologiesused for designing analogue integrated circuits are bipo-
lar and MOS. Continuous improvement of analogue MOS capabilities forced the designers
to explore the world of CMOS technology. CMOS stands for the complementary MOS and
this technology makes use of both p-channel and n-channel MOSFETs.
In standard 0.18 µm CMOS technology, the epitaxial substrate has low bulk resistivity
of the order of 10−2Ω·cm. 0.18 µm process provides a single polysilicon layer and six metal
layers. Metal layers are fabricated using aluminium and numbered from M1 to M6. Top
layers M5 and M6 have a thickness of 0.92 µm while lower layers have typical thickness
4.3 10 Gbit/s CMOS Differential Amplifier 69
of 0.5µm, respectively. The metal layers in CMOS are separated by the internal dielectric
material having dielectric constant εr≈4.3. For substrate and p-well isolation, the deep
n-well isolation (NISO) technology is available.
Both n- and p-channel MOSFETs are fabricated as surface devices. The gates are fab-
ricated with n- or p-type polysilicon layer and topped with a metal silicide for lower gate
series resistance. Figure 4.13 shows the cross-section of 0.18 µm CMOS transistors and
transconductance characteristics of n-channel MOSFET as a function of the gate-source
voltage.
p-well
n+n+n+
p+
NISO
n-MOS
transistor
n-well
p+n+
p+
p-MOS
transistor
p+ substrate
p+ epitaxial
0
0.1
0.2
0.3
0.4
0.5
0.2 0.4 0.6 0.8 11.2
Transconductance
[mS/µm]
Vgs[V]
Vgs= Vds,
w/o body effect
Figure 4.13: Cross-section of the 0.18 µm CMOS process (left) and n-MOS transconductance as a function
of the gate-source voltage (right)
4.3.1 Design of Transmission Line Structures
Recent advances in high-speed circuits have highlighted the interest in monolithic transmis-
sion lines – as both parasitic components and useful devices. With operating frequencies
increasing to several tens of GigaHertz and/or the chip dimensions approaching several
millimeters, the transmission lines can degrade the performance of analogue and digital
circuits. Therefore, accurate modelling of transmission lines is thus necessary in the design
and analysis of high-speed circuits. In circuit design, several characteristic properties of
transmission lines become critical: characteristic impedance, loss, wave velocity, and field
confinement [101].
Three main types of transmission line structures that can be used in CMOS design are
the Microstrip lines (MS), Coplanar Waveguides (CPW), and Striplines (SL). The next
subsection presents the simulation results on microstriplines, coplanar waveguides and
striplines using the Momentum Electromagnetic Field Simulator and LineCalc in ADS.
50 Ωlines are fabricated in 0.18 µm CMOS process. The transmission line models are
extracted from measured S parameter data. Later on, these models are used to design the
differential amplifier using striplines in 0.18 µm CMOS technology.
70 Chapter 4 High-Speed Integrated Circuits for oDPSK Transmission
Microstrip Line
Figure 4.14 (left) shows a geometry of the microstripline in CMOS process. As the thick-
ness of the dielectric layer depends on the CMOS process, the characteristic impedance of
microstriplines is controlled by the width (W) of the conductor ribbon fabricated in top
metal M6 which is deposed on the internal CMOS dielectric with ground metallization in
bottom metal M1 to prevent any interaction with lossy substrate. The layers are chosen
such that capacitance per unit length is minimized, while the width of signal conductor is
maximized. Thus, the smallest possible attenuation is obtained.
Varying the conductor width from 29 µm to 3µm, changes the characteristic impedance
of the microstripline in a range of 30-100 Ωas shown in Figure 4.14 (right). A 50 Ω
microstripline line is fabricated using 13 µm width.
H
M6
M1
T
0
20
40
60
80
100
120
010 20 30
Width ???m ?
Zo ( Ω )
measured
Width [µm]
Zo[Ω]
Figure 4.14: Geometry of the microstripline in CMOS (left) and its characteristic impedance as function of
the conductor width (right)
Coplanar Waveguide
Figure 4.16 shows the geometry of the coplanar waveguides using the top metallization
layer M6 in CMOS. The characteristic impedance of coplanar waveguide is controlled by
the width (W) of the central conductor and the gap (G) between the central conductor and
ground planes.
M6
Figure 4.15: Geometry of coplanar waveguide
4.3 10 Gbit/s CMOS Differential Amplifier 71
Simulated variation of the characteristic impedance for coplanar waveguide as a func-
tion of aspect ratio W/(2G+W)is given in Figure 4.16 (left). The characteristic im-
pedance of the CPW is in the range of 40 −120 Ω while keeping the W+ 2G= 39.5µm
constant, as shown in Figure 4.16 (right). Two slots with the same width G=4.75 µm, sep-
arated by a 30 µm metallic ribbon give 50 Ωwaveguide, obtained for minimum total loss
in a frequency range up to 10 GHz. Due to the stronger electromagnetic coupling to the
lossy substrate coplanar waveguide exhibits higher loss at higher frequency than microstrip
line. Figure 4.17 shows the measured attenuation results in dB/mm from the test struc-
tures. Wider signal line is more lossy at higher frequency duo to the higher substrate loss,
indicated by the large width (W) of the signal line and the large gap G.
20
40
60
80
100
120
140
0 10 20 30 40
Width [µm]
2G+W=39.5
measured
20
40
60
80
100
0,2 0,4 0,6 0,8 1
W/(2G+W)
Zc[Ω]
Z
c
[
Ω
]
Figure 4.16: Characteristic impedance of the coplanar waveguide in function of the ratio W/(W+2G)(left)
and conductor width (right)
0
0.4
0.8
1.2
1.6
2
15 25 35
Width (µm)
Attenuation [dB\mm]
10GHz
20GHz
40GHz
Width [µm]
Figure 4.17: Measured attenuation for 50 ΩCPW versus width of signal line
Stripline
Figure 4.18 (left) shows the geometry of the stripline (SL) which is placed in the metal
M4 of the CMOS process. Normally, the metal M4 is sandwiched between the two parallel
metal planes, bottom metal (M1) and top metal (M6). Striplines are typically realized using
narrow width due to the fact that the two ground planes which are in the close vicinity of
the stripline introduces the large parasitic capacitance. 3.5µm wide stripline brings the
characteristic impedance close to 50 Ω. Changing the conductor width from 12 −2µm,
changes the characteristic impedance of the stripline in a range of 20 −60 Ω as shown in
Figure 4.18 (right).
72 Chapter 4 High-Speed Integrated Circuits for oDPSK Transmission
H
M1
M6
M4Tεr
0
20
40
60
80
036912 15
Width(?m)
Zo(W)
Width [µm]
Zo[Ω]
measured
Figure 4.18: SL configuration (left) and characteristic impedance of the SL in function of the conductor
width (right)
Characterization of Transmission Line Structures
Figure 4.19 shows the microphotograph of the interconnect test structures (MS, CPW, and
SL) used to extract the transmission line parameters. The striplines are invisible as they are
fabricated in an intermediate metal layer M4. Interconnect test structures also includes the
open structures for pad calibration which has same layout as the interconnect test structures
but without the central conductor (not shown). The transmission line structures are charac-
terized by S-parameter measurement in the frequency domain in range from 100 MHz to
40 GHz.
Figure 4.19: Microphotograph of the fabricated MS, CPW and SL (from left to right)
Figure 4.20 shows the measured losses for microstripline, coplanar waveguides, and
striplines. Attenuation at 10 GHz is 0.45 dB/mm and 0.4 dB/mm for microstripline and
coplanar waveguide, respectively. Due to the narrower width of the stripline, the attenu-
ation is comparatively larger with respect to microstripline and coplanar waveguide. It is
typically around 1.75 dB/mm at 10 GHz.
Frequency variation of the interconnect test structures are extracted directly from S–
parameter measurement data. This information is essential for developing accurate and
verified transmission line models for interconnect test structures. The simulation models up
to 20 GHz are developed using the classical Telegraph model [102]. The detailed procedure
is described in Appendix C. Table 4.1 lists the distributed circuit parameters (R,L, and C)
extracted for per millimeter length of microstriplines, coplanar waveguides, and striplines.
4.3 10 Gbit/s CMOS Differential Amplifier 73
5 10 15 20 25 30 350 40
-2.5
-1.5
-0.5
0.5
-3.5
Frequency [GHz]
S21[dB]
5 10 15 20 25 30 350 40
-30
-25
-20
-15
-35
-10
Frequency [GHz]
MS (O), CPW () and SL (∆)
S11[dB]
MS (O), CPW () and SL (∆)
Figure 4.20: Comparison of measured MS, CPW and SL data for the magnitude S21 and S11
Table 4.1: Distributed circuit parameters for interconnect test structures
Type of TL R(Ω/mm) L(nH/mm) C(pF/mm)
MS 5.2 0.32 0.13
CPW 5.0 0.28 0.17
SL 22.5 0.30 0.19
Extracteddistributedresistance Rof SL is almostfour times largerthan the resistanceof
MS and CPW. As a result, the attenuation of SL is also larger as compared to the attenuation
of MS and CPW. Figures 4.21, 4.22 and 4.23 show the comparison of measured, simulated
and modelled S21 magnitude and S21 phase data for MS, CPW and SL, respectively.
510 15020
-1.0
-0.5
0.0
-1.5
0.5
freq, GHz
S21(dB)
510 15020
-40
-20
-60
0
freq, GHz
S21(dB)
S21(dB)
Modelled
Measured
Simulated
Modelled
Measured
Simulated
Frequency [GHz]
Phase S21 [°]
S21[dB]
Frequency [GHz]
Figure 4.21: Comparison of measured, simulated and modelled MS data for the magnitude S21 (top-left),
phase S21 (top-right) and magnitude S11 (bottom)
74 Chapter 4 High-Speed Integrated Circuits for oDPSK Transmission
510 15020
-1.0
-0.5
0.0
-1.5
0.5
freq, GHz
S21(dB)
510 15020
-40
-20
-60
0
freq, GHz
S21(dB)
Modelled
Measured
Simulated
Modelled
Measured
Simulated
S21[dB]
Frequency [GHz]Frequency [GHz]
Phase S21[°]
Figure 4.22: Comparison of measured, simulated and modelled CPW data for the magnitude S21 (top-left),
phase S21 (top-right) and magnitude S11 (bottom)
Modelled
Measured
Simulated
510 15020
-2.0
-1.5
-2.5
-1.0
freq, GHz
S21(dB)
510 15020
-40
-20
-60
0
freq, GHz
S21(dB)
Modelled
Measured
Simulated
S21[dB]
Frequency [GHz]
Phase S21[°]
Frequency [GHz]
Figure 4.23: Comparison of measured, simulated and modelled SL data for the magnitude S21 (top-left),
phase S21 (top-right) and magnitude S11 (bottom)
4.3.2 Circuit Design
Recently, CMOS has emerged as an alternative to compound semiconductor or SiGe bipo-
lar technology for integration of microwave and optical front-end circuits due to improve-
ment of the RF performance, integration capability with baseband circuits, and most impor-
tantly the low cost. The motivation behind this work is to explore the limitation of CMOS
distributed amplifier circuit design and analysis together with CMOS device and passive
element modelling.
Equation (4.5), described in section 4.1, reduces to N= 1 where Nis the optimum
number of stages required to maximize the gain of the CMOS distributed amplifier using
striplines at 10 GHz and/or beyond. This indicates that the distributed gain using striplines
is not possible in CMOS technology. However, as the striplines have a certain advantage in
complex analogue circuitry for system integration, to study their behavior and limitations at
10 Gbit/s data rate, a differential amplifier using striplines was realized in 0.18 µm CMOS
technology. The amplifier consists of one differential stage followed by three pairs of
distributed common-source common-gate (cascode) stages.
4.3 10 Gbit/s CMOS Differential Amplifier 75
Vss
VIN1
I0
vIN2
Vb
Vb
Vb
Vb
Vd
Vd
VbVb
Out
Out
Vd
Vd
Figure 4.24: Schematic of the differential amplifier using striplines in CMOS
Figure 4.24 shows the schematic circuit diagram of the differential amplifier. Differ-
ential input signals are dc coupled to the differential preamplifier. The external gate bias
of 1.3 V is applied through the bias Ts. The preamplifier input stage has on-chip 50 Ωre-
sistors to provide good impedance matching. The two output signals from the differential
preamplifying stage are used to drive common-source common-gate stages which operate
from single positive DC power supply of 1.8 V. A total length of the stripline’s in each
branch of the differential amplifier is 665 µm.
Figure 4.25: Microphotograph of the realized chip
Figure 4.25 shows a microphotograph of the fabricated amplifier. The circuit elements
are arranged symmetrically in layout to minimize the offset. Decoupling capacitors are
connected between the power supply voltages and the ground. The power dissipation is
185 mW on a chip having an active area of 0.5 mm2.
76 Chapter 4 High-Speed Integrated Circuits for oDPSK Transmission
Figure 4.26 shows the simulated (differential input and single-phase input) and mea-
sured (single-phase input) S21-parameter as a function of frequency. The simulated gain
is 11.5dB over the 7 GHz frequency range. The characteristic frequency response of the
amplifier is measured using the wafer probe workstation and a HP network analyzer. To
perform this measurement, the another differential input and the corresponding differen-
tial output is terminated with a 50 Ωresistor. The measured gain and 3 dB bandwidth is
6dB and 6.2 GHz respectively. The simulated frequency response includes the effect of
striplines but not the effect of parasitic layout capacitances. Thus, there is a slight mismatch
of simulated and measured S21-parameter.
-5
0
5
10
15
1.0E+08 1.0E+09 1.0E+10 1.0E+11
Simulated differential
Measured single phase
Simulated single phase
Frequency [Hz]
S21 [dB]
Figure 4.26: Comparison of measured, simulated single phase and simulated differential magnitude of S21
Measured input and output reflection coefficients show acceptable performance. S11 is
typicallybelow-9.5dBand S22 is below -7dB upto 10GHzrange, as shown inFigure 4.27.
-40
-30
-20
-10
0
0,0E+00 5,0E+09 1,0E+10 1,5E+10 2,0E+10
S22(dB)
-40
-30
-20
-10
0
0,0E+00 5,0E+09 1,0E+10 1,5E+10 2,0E+10
Measured single phase
Simulated single phase
Frequency [Hz]
S11 [dB]
Frequency [Hz]
S22 [dB]
Measured single phase
Simulated single phase
Figure 4.27: Comparison of measured, simulated single phase and simulated differential magnitude of S11
Figure 4.28 shows the exemplary measured eye diagram in single-phase configuration
using a 10 Gbit/s 27−1pseudorandom bit sequence signal (PRBS) having 50 mVpp input
amplitude. The differential structure also rejects the common mode interferences by 8 to
16 dB over 10 GHz frequency range.
4.4 Conclusion 77
-100 Voltage [mV]
100
Time [ps], 50ps per unit
Figure 4.28: Measured eye diagram at 10 Gbit/s for 27−1PRBS input signal
4.3.3 Result Disscusion
Simulation models for the MS, CPW and SL are extracted to give fundamental insight in
to transmission line structures realized in 0.18 µm CMOS technology. A differential am-
plifier was designed and simulated using the above developed transmission line models.
The differential amplifier was later fabricated in 0.18 µm CMOS technology. The experi-
ment demonstrates 10 Gbit/s signal propagation over narrow CMOS striplines. This opens
the possibility of using striplines whenever over all good shielding is needed in complex
analogue circuits.
The differential amplifier has a 3 dB bandwidth of 6.2 GHz. For single-phase input,
the amplifier has the voltage gain of 6 dB at 10 Gbit/s and a CMRR of 8 dB. This is
due to the fact that MOS transistor used in the constant current source of the differential
amplifier has a very large size and is being operated with minimum gate-to-source voltage
in order to provide larger bandwidth. To increase the CMRR of the amplifier, it is necessary
to reduce the size of the current source MOS transistor as well as to provide the large
voltage headroom. For n-channel MOS transistor, this is only possible by using triple-well
technology [103]. The idea is to place the n-channel devices in p-wells, which are isolated
from each other by n-wells and a third buried n-isolation implant. Simulation results show
that two times better CMRR can be obtained by using this technology.
This CMOS circuit using striplines exhibits comparable performance with that of the
state-of-the-art amplifiers designed in conventional technologies [44–46]. However, the
results published in [45] and other similar publications have higher gain due to the fact
that they have a significant transimpedance gain with very similar performance bandwidth.
None of the referenced publication reports on the CMRR performance of the amplifier.
4.4 Conclusion
This chapter presents the simulation results on the design of differential travelling wave
amplifier in pseudomorphic AlGaAs/InGaAs HEMT technology which combines the ad-
vantages of a differential and traveling wave designs. The circuit has 17 dB of differential
78 Chapter 4 High-Speed Integrated Circuits for oDPSK Transmission
gain, 3 dB bandwidth of 46 GHz and a CMRR of 16.5 dB. The circuit could be employed
as a differential linear amplifier for the reception of RZ-DPSK signals using the balanced
optical front end to obtain high receiver sensitivity and a better OSNR.
Simulation models for the microstrip lines, coplanar waveguides and striplines are ex-
tracted from the measured S-parameter data. Simulated and measured results on the inter-
connect transmission line structure are compared. Moreover, the accurate models are later
developed for advance simulation.
Finally, a 10 Gbit/s differential in differential out linear amplifier using striplines is
designed and fabricated in 0.18 µm CMOS technology. The experiment demonstrates a
voltage gain of 6 dB with CMRR of 8 dB with a good eye opening at 10 Gbit/s.
Chapter 5
Result Discussion and Future Scope
The state of the art 40 Gbit/s differential phase shift keying and differential quadrature
phase shift keying system was in-house developed using commercially available compo-
nents. It was shown that signed online chromatic dispersion detection technique also works
satisfactorily for the above mentioned modulation formats. The same technique was also
used to adaptively compensate the chromatic dispersion of the various transmission spans
using a thermally tunable dispersion compensator for RZ-DPSK modulation format. Fur-
thermore, the DQPSK spectral efficiency was once more doubled using polarization divi-
sion multiplexing and a 160 Gbit/s transmission per WDM channel was demonstrated using
a thermally tunable dispersion compensator.
In order to implement optical transmission systems using either DPSK or DQPSK mod-
ulation formats, some special high–speed integrated circuits (ICs) are required. Therefore,
a differential amplifier employing distributed amplifying stages that could be used to am-
plify the signals coming out of the balanced 40 Gbit/s DPSK/DQPSK optical receivers was
simulated and designed in a pseudomorphic AlGaAs/InGaAs HEMT technology. Also a
differential amplifier using striplines at 10 Gbit/s was designed, simulated, and later on
fabricated in 0.18 µm CMOS technology.
5.1 DPSK Transmission
Return-to-zero differential phase shift keying transmitter at 40 Gbit/s is developed in-house
under the supervision of R. No`
e. The optical transmitter uses a DFB laser at 192.5 THz
as a source while a dual-drive Mach-Zehnder modulator was used to generate nonreturn-
to-zero differential phase shift keying signal (NRZ-DPSK). Another dual-drive modulator
was used as a pulse carver to generate Return-to-zero differential phase shift keying signal
(RZ-DPSK).
Similarly, in-house developed balanced optical receiver uses a Mach-Zehnder inter-
ferometer having 4-bit delay as a delay demodulator and two high-speed photodiodes for
balanced detection. This delay interferometer was thermally stabilized using the home
made temperature controller. It was tuned using the differential micro-heaters with total
79
80 Chapter 5 Result Discussion and Future Scope
constant power. Main problem associated with this photonic lightwave circuit (PLC) is that
it had nonnegligible polarization-dependent loss. As a result, the balanced optical receiver
suffers nonnegligible transmission penalty due to polarization dependent phase shift that
exist in the interferometer. One possibility to resolve this issue is to automatically adjust its
phase using the available differential micro-heaters. This kind of automation can be imple-
mented using the lock-in detection scheme. Therefore, one of the photodiode output was
tapped using a pick-off T. The signal at the pick-off T was amplified using a chain of five
10 dB amplifiers at 10 GHz followed by a RF power detector. The output of the RF power
detector drives the 400 Hz analogue lock-in amplifier and its output was used to stabilize
the receiver interferometer. This scheme eliminates the polarization dependent phase shift
and makes the system more robust with respect to temperature and occurring polarization
changes in the fiber spans, and also to the laser frequency offsets [59].
The signed online chromatic dispersion (CD) detection scheme, previously demon-
strated by D. Sandel for NRZ signals, was implemented, for the first time, for DPSK
modulation format. This scheme introduces hardly any transmission penalty for DPSK
modulation format using a balanced optical receiver even with an interferometer having
4-bit delay. The CD detection scheme offers the several advantages such as low cost of
integration and high sensitivity of around 1ps/nm in 1 ms measurement interval. It has
wide dynamic range and detects the sign of the CD without any ambiguity.
Moreover, the residual chromatic dispersion of a 263 km fiber link at 40 Gbit/s for
NRZ-DPSK and CS(RZ)-DPSK was compensated, for the first time, using the automatic
control of a thermally tunable fiber Bragg grating based dispersion compensator. The to-
tal measured penalty of transmission and CD compensation was -1.2 dB ... +1.2 dB, for
various link lengths and compensator CDs. As evident from our experiments, the CS(RZ)-
DPSK modulation format outperforms the conventional modulation formats and hence we
recommend it for long and ultra-long haul optical fiber transmission.
5.2 DQPSK Transmission
Spectral efficiency of the (CS)RZ-DPSK modulation format was doubled using differential
quadrature phase shift keying (DQPSK) signal. The NRZ-DQPSK signal was generated
interferometrically, using NRZ-DPSK signals. The transmitter interferometer was stabi-
lized using a 10 KHz lock-in detection scheme. This 10 KHz phase modulation has a depth
of 0.01 rad (rms) and does not introduce any transmission penalty. Carrier suppressed
return-to-zero (CSRZ)-DQPSK signal was generated using a subsequent dual-drive Mach-
Zehnder modulator driven at half of the clock rate.
2×40 Gbit/s RZ-DQPSK transmission over 263 km fiber link with back-to-back re-
ceiver sensitivity of −27.5dBm and Q factor >20 dB is demonstrated [30]. The exper-
iment shows sufficient resilience against non-linear phase noise and band limitation in a
40 Gbit/s WDM DEMUX with Q=17.5 dB. Residual chromatic dispersion was manually
tuned using a thermally tunable dispersion compensator. Especially, for this modulation
format the automatic control of the tunable dispersion compensator using signed online
5.3 High-Speed Integrated Circuit for oDPSK Transmission 81
CD detection technique was not possible due to the nonnegligible polarization dependent
loss of the receiver interferometer. It is very difficult to stabilize the receiver interferometer
for DQSPK modulation format using the lock-in stabilization technique unless the receiver
interferometer is free from the polarization dependent loss. Future scope of this work would
be to either realize the delay interferometer with negligible polarization dependent loss or
to implement automatic polarization control using the lithium niobate-based polarization
transformers for the interferometer Other possibility is to investigate or develop a new kind
of lock-in stabilization technique that may work even in the presence of polarization de-
pendent loss.
Further more, the spectral efficiency of the DQPSK signal was once more doubled us-
ing polarization division multiplex transmission [75]. This 160 Gbit/s transmission per
WDM channel was demonstrated over 230 km of fiber with Q factor >15 dB using the
automatic polarization control and a manually tunable dispersion compensation. A future
scope of this work would be to transmit 160 Gbit/s signals with 40 DWDM channels having
100 GHz channel spacing in one single band in order to demonstrate the Tbit/s transmission
capacity and spectral efficiencies beyond 1 bit/s/Hz. Fortunately, to date, such multi-terabit
transmission experiments are recently demonstrated by [73,74]. Future scope of this would
be to transmit 40 DWDM channels with fully automatic tunable chromatic dispersion com-
pensation using low-cost signed online chromatic dispersion detection scheme.
5.3 High-Speed Integrated Circuit for oDPSK Transmis-
sion
A differential amplifier using distributed stages is designed and simulated in a pseudomor-
phic AlGaAs/InGaAs HEMT technology [35]. The circuit combines the two different con-
cepts: differential and travelling wave concept, achieving the simulated flat gain of around
17 dB over the 46 GHz frequency range with a CMRR of 16.5 dB. Layout of this circuit
was done using Cadence software. However, the circuit could not be fabricated due to sev-
eral reasons. Future scope of this work would be to fabricate and characterize the circuit
and to study and compare its measured characteristic properties with the simulated ones.
Nevertheless, the latest publication [86] shows that the record sensitivity (39 photons/bit)
can be achieved for the RZ-DPSK modulation format using differential travelling wave
amplifier having a CMRR of >20 dB. Therefore, the further step would be to resimulate
and redesign the differential distributed amplifier for higher CMRR using the experience
gained using these trials.
A single-stage differential pre-amplifier followed by three pairs of spatially distributed
common source stages using striplines, was designed and fabricated in standard 0.18µm
CMOS technology. For a single-phase input, amplifier has a gain of 6dB at 10 Gbit/s.
The measured 3 dB bandwidth is 6.2 GHz. The CMRR at 10 Gbit/s is 8 dB. The ex-
periment demonstrates the possibility of using the striplines in low-cost standard CMOS
technology at 10 Gbit/s. Amplifier performance is comparable with state-of-the-art ampli-
82 Chapter 5 Result Discussion and Future Scope
fiers designed in conventional technologies. This experience further open the possibilities
of using striplines in CMOS technology where over all good shielding is needed. Further
scope of this work would be to improve the CMRR of the amplifier as has been already
discussed in chapter 4.
5.4 Conclusion
This chapter summarizes transmission experiments done at 40 and 160 Gbit/s using the
latest modulation formats such as return-to-zero–differential phase shift keying and differ-
ential quadrature phase shift keying which are currently under investigation for their future
use in digital optical transmission systems. It also discusses the simulation and measure-
ment results obtained on the differential amplifiers at 10 Gbit/s and 40 Gbit/s.
Appendix A
Definitions
A.1 Bit Error Rate
The performance of a digital optical communication system is characterized by the bit-
error-rate (BER). BER is defined as the average probability of incorrect bit identification.
Therefore, a BER of 10−6corresponds to, on average, one error per million bits. Most
of the digital optical communication systems specify a BER of ≤10−9. The BER with
optimum setting of the decision threshold is given by
BER = 1
2erfc( Q
√2)(A.1)
where erfc stands for the complementary error function and Q-factor is given by
Q=µ1−µ0
σ1+σ0
.(A.2)
In the above expression for Q-factor, the µ1and µ0are the average intensity levels of the
transmitted “1” and “0”, while σ1and σ0are standard deviation of the noise sources asso-
ciated with the transmitted “1” and “0”, respectively. The expression (A.1) is not accurate
enough since the derivation is based on the Gaussian approximation for the receiver noise
statistics. The approximate form of ( A.1) obtained by using the asymptotic expansion of
erfc(Q/√2) is reasonably accurate for Q > 3and is given by
BER ≈exp(−Q2/2)
Q√2π.(A.3)
The BER improves as Q-factor increases and becomes lower than 10−12 for Q > 7. The
receiver sensitivity corresponds to the average optical power for which Q= 6, since
BER ≈10−9when Q= 6.
For the asynchronous ASK case,
BER ≈1
2exp(−ηNp/4) (A.4)
83
84 Appendix A Definitions
where ηis the quantum efficiency of the photodetector and Npis a number of photons re-
ceiving during the bit “1”. If we assume η= 1 the above equation shows that BER = 10−9
for Np= 80. For ASK, this method gives a fairly good prediction of the BER, although
the noise distribution in the intensity domain is not exactly Gaussian [56]. However, direct
use of ( A.4) in DPSK may lead to wrong prediction of the BER even in the linear regime
due to the fundamentally non-Gaussian nature of the noise distribution in the output signal
of the DPSK balanced receiver. The BER calculation is slightly more complicated for the
DPSK case since is formed by the difference of two photo-currents. The final result is,
however, quite simple and is given by
BER ≈1
2exp(−ηNp).(A.5)
For η= 1, a BER of 10−9is obtained for Np= 20. Thus DPSK provides 3 dB more system
margin, than ASK.
A.2 Receiver sensitivity
An important parameter that indicates the receiver performance is called receiver sensitiv-
ity. It is defined as minimum average received optical power for which the BER of the
optical receiver is 10−9. The receiver sensitivity depends on the optical signal-to-noise ra-
tio, as defined later, which in turn depends on various noise sources that corrupt the received
optical signal.
A.3 Optical signal-to-noise ratio
The optical signal-to-noise ratio (OSNR) is the measure of the ratio of signal power to noise
power in an optical channel. It is usually measured with an Optical Spectrum Analyzer
(OSA) and defined as
OSNR = 10 log S
N(A.6)
where the symbol S represents the (linear) optical signal power and the symbol N is the
(linear) optical phase power.
OSNR is important because it suggest a degree of impairment when the optical signal
is carried by an optical transmission system that includes optical amplifiers. The detection
of the signal is typically affected by attenuation and dispersion. With the use of the ampli-
fiers, there is the additional impairment because the of noise seen in the receiver due to the
presence of Amplified Spontaneous Emission noise (ASE). In practice, the use of an am-
plifier will help improve the signal because the increase in the signal amplitude will help
overcome noise generated in the receiver’s front end. However, the optical background
(noise) that accompanies the desired optical signal will be amplified along with the signal;
consequently the OSNR will tend to degrade as it passes trough the transmission system.
A.3 Optical signal-to-noise ratio 85
The optical noise near signal wavelength can impair the receiver ability to properly
decode the signal because of optical interference between the optical signal and optical
noise. This impairment can be a bigger contributor to the BER than the power fluctuation
in the optical noise.
86 Appendix A Definitions
Appendix B
Theory of the Traveling Wave Amplifier
The basic idea behind the design of traveling wave amplifier is to absorb the parasitic
capacitances of the transistors into the transmission line and making them as a part of the
passive network. For this explanation, there is a need to develop the expression for the
complex propagation constant, γ, of a lossy transmission line shown in Figure B.1.
C
L
CG G
L L
Figure B.1: Lumped transmission line with shunt loss
Generally,
γ=α+jβ =p(R+jωL)(G+jωC).(B.1)
In this expression and in Figure B.1 the elements R,L,Gand Cmay be expressed in per
section terms for a lumped component artificial transmission line, or in per unit distance
terms for a uniform transmission line. αand βare the attenuation and phase constants,
respectively. The characteristic impedance, Z, of the transmission line is
Z=sR+ωL
G+ωC .(B.2)
Combining above two equations gives
γ= (G+ωC)Z. (B.3)
For G << ωC and R << ωL,γmay be approximated as follows:
γ= (G+ωC)ωL
G+ωC 1/2
(B.4)
87
88 Appendix B Theory of the Traveling Wave Amplifier
The further simplification of (B.4) leads to:
γ= (G+ωC)L
C1/21 + G
ωC −1/2
(B.5)
where for G << ωC, the expression (1 + G/ωC)−1/2can be approximated by the first
order of McLaurin series as 1−G/2ωC, and this leads to
γ=G+ωC −G
2−G2
2ωC rL
C.(B.6)
For G << ωC and (G2/2ωC)pL/C ≈0the expression (B.5) can be written as:
γ=G
2+ωCrL
C(B.7)
Thus, the attenuation constant, α, and the phase constant, β, are given by
α≈G
2rL
C
β≈ω√LC
.(B.8)
For lossless case, Z0even can be approximated by
Z0≈rL
C.(B.9)
In a practical amplifier the inductors are realized with the lengths of transmission line.
Regardless, the inductors not only have inductance per unit length but they also have capac-
itance associated with them. This capacitance must be included together with the device
capacitance in full analysis.
For the drain line, the transmission line section between the FETs are taken to be of
electrical length ld. The total effective inductance per unit length of the transmission line
will be termed L0
d. If the capacitance per unit length of the transmission line is C0
d, the total
capacitance, C0
D, per unit length of the drain line is given by
C0
D=C0
d+Co
ld
(B.10)
where Cois the total output capacitance of the transistor. The Figure B.2 shows a section
of the drain line, from which the shunt conductance per unit length, GD
GD= (1/r0)×1/ld(B.11)
Using the results of equations (B.8) and (B.9), the attenuation per unit length along the
89
CoCd' ld
2
Cd' ld
2
Ld' ld
FET
ld
ro
Figure B.2: One section of the drain line
drain line is
αD=1
2r0ldsL0
d
C0
d+Co/ld
.(B.12)
The phase constant in radians per unit distance along the drain line is
βD≈ωqL0
d(C0
d+Co/ld).(B.13)
The characteristic impedance is
Z0D≈sL0
d
C0
d+Co/ld
.(B.14)
Cgs
Cg' lg
2
Cg' lg
2
Lg' lg
FET
lg
R
Figure B.3: One section of the gate line
For the gate line the procedure is almost same as for the drain line except for the fact
that the equivalent circuit of the transistor is now a series RC network. The impedance of
the series RC network is given by
Z=R+1
jωCgs
(B.15)
The admittance is
Y=Z−1=jωCgs +ω2C2
gsR
1 + ω2C2
gsR2(B.16)
90 Appendix B Theory of the Traveling Wave Amplifier
In the above expression, ω2C2
gsR2<< 1so that one may take the denominator roughly
equal to unity and hence
Y≈jωCgs +ω2C2
gsR. (B.17)
Following a procedure identical to that employed for the drain line, we can now write,
for the gate line, the inductance per unit length is given by L0
g. Similar to drain line the C0
g
is the capacitance per unit length of the gate transmission line, C0
Gis the total capacitance
per unit length of the gate line, Cgs is the capacitance of the transistor.
C0
G=C0
g+Cgs
lg
(B.18)
The conductance per length is given with
G0
G=ω2C2
gsR
lg
(B.19)
Making a use of equations (B.8) and (B.9), the attenuation and phase constant, αGand βG,
of a gate line are expressed as
αG≈ω2C2
gs R
2lgsL0
g
C0
g+Cgs/lg
(B.20)
βG≈ωqL0
g(C0
g+Cgs/lg)(B.21)
and gate impedance is given by
Z0G≈sL0
g
C0
g+Cgs/lg
.(B.22)
Equations (B.12) and (B.20) respectively give the attenuation coefficient of signals
propagating along the drain and gate lines. Similarly, equations (B.13) and (B.21) give
the phase delay per unit length along the two lines and equations (B.14) and (B.22) give
their characteristic impedances. All this quantities appear in the expression for the power
gain of the amplifier which will now be derived.
Consider the path from the input, via FET k, to the output, as shown in Figure B.4. The
signal wave propagating along the gate line of FET khas an amplitude
Vgk =Vgo exp(−αGklg).(B.23)
FET kgenerates a current Idk in response. Assuming half of Idk goes into the wave prop-
agating towards the external load, the current actually entering the load has an amplitude
of
Iok =Idk
2exp[−αD(N−k)ld](B.24)
91
But
Idk =gkoVgk =gkoVgo exp(−αGklg)(B.25)
Therefore
Iok =gko
2Vgo exp(−αDNld) exp[k(αDld−αGlg)].(B.26)
Iok
k lg
Idk
Vgk
(N-k) ld
k-th HEMT
ZOD
Vgo
Figure B.4: Signal path from the input to the output via the ’k’-th transistor
If the phase delay per section of the gate line is equal to the phase delay per section
of the drain line, all components Iok arising from the individual FETs will add in-phase so
that the total current in the external load, Iout, can be found by summing the magnitudes
Iok given by equation (B.26). The sum is taken over 1 ≤k≤N. This summation can be
replaced by an integration. Taking the integration limits from 1to Nwould be appropriate
if amplifier uses Nnumber of FETs. But taking the limits from 0to Nis even more
appropriate given that the total length of the lines involved and hence for determining the
total attenuation. Since the integration affects the last term in equation (B.26), a term which
expresses attenuation assume limits from 0to N. Thus, the total output current is
Iout =gko
2Vgo exp(−αDNld)ZN
0
exp[k(αDld−αGlg)]dk
=gko
2Vgo
exp(−αGNlg)−exp(−αDNld)
αDld−αGlg
(B.27)
The amplifier power gain is defined as
Gp=Pout
Pin
.(B.28)
The input power to the amplifier is given by
Pin =V2
go
ZOG
(B.29)
92 Appendix B Theory of the Traveling Wave Amplifier
while the amplifier output power is given by
Pout =I2
outZOD.(B.30)
As a result, the power gain is given by the following equation which is used in chapter 4.
Gp=g2
koZOGZOD[exp(−αGNlg)−exp(−αDNld)]2
4(αDld−αGlg)2(B.31)
Appendix C
Extraction of Transmission Line
Parameters
Interconnect signal transmission is based on the solution of the classical Telegraph trans-
mission line equation [102]. A models has been developed that presents, in the frequency
domain the interconnect voltage and current in terms of propagation constant, γand cat-
achrestic impedance, Z. Then the impedance and the propagation constant are described
by four four distributed transmission line parameters, R,L,C, and G. An infinitely small
section of this model incorporating distributed circuit elements are shown in Figure C.1.
Figure C.1: Single transmission line represented by a two-port network and described with distributed trans-
mission line parameters R, L, C and G→0
These distributed circuit parameters describe per unit length values and not the lumped
element values that are assumed in simulation. IC interconnect circuit elements Z,γ,R,
L,C, and Gare shown to be function of frequency. To solve the Telegraph equation for IC
interconnect- signal propagation requires an understanding of S-parameter based intercon-
nect test structure characterization. Single line IC interconnect transmission i s represent
by a two port network and is tested in a controlled (Zo= 50Ω) impedance microwave
measurement system. The S-parameter response measured from a lossy unmatched trans-
mission line with parameters γand Zin a Zoimpedance system are:
S=1
Ds(Z2−Z2
0) sinh γl 2ZZ0
2ZZ0(Z2−Z2
0) sinh γl(C.1)
93
94 Appendix C Extraction of Transmission Line Parameters
where
Ds= 2ZZ0cosh γl + ( Z2+Z2
0) sinh γl. (C.2)
Since the above matrix is symmetrical it contains two independent linear equations.
This S-parameter matrix is converted to ABCD parameters, which incorporated the inter-
connect propagation constant γ(ω)and impedance Z(ω)more explicitly. The equivalent
ABCD matrix is
ABCD=cosh γl Z sinh γl
(1/Z) sinh γl cosh γl (C.3)
The relationship between S parameters and ABCD matrix is:
A=(1 + S11 −S22 −4S)
2S21
(C.4)
B=(1 + S11 +S22 +4S)Z0
2S21
(C.5)
C=(1 −S11 −S22 +4S)Z0
2S21Z0
(C.6)
D=(1 −S11 +S22 −4S)
2S21
(C.7)
where
4S=S11S22 −S21S12 (C.8)
The equations (C.1)– (C.7) are combined to yield
e−γl =1−S11 2+S21 2
2S21 ±K−1
(C.9)
where
K=(S11 2−S21 2+ 1)2−(2S11)2
2S21 21/2
(C.10)
and
Z2=Z02(1 + S11)2−S21 2
(1 −S11)2−S21 2(C.11)
During the extraction of complex parameter γand Zfrom e−γand Z2, the cylindrically
mapped phase output of the S-parameter network analyzer (−180◦to 180◦) is converted to
the true radian measurement phase which can be any real value. Also, extracted parame-
ter with values that are not physical satisfy the Telegraph equation for propagation in the
95
negative direction. Once γand Zare determined, then from standard transmission line re-
lationships, characteristic impedance, Z, (B.1) and complex propagation constant, γ, (B.2),
given in Appendix B, follows
R=Re{γZ}
L=Im{γZ}/ω
G=Re{γ/Z}
C=Im{γ/Z}/ω
(C.12)
Thus, the Telegraph equation transmission line model parameters are determined by
combining (C.12).
96 Appendix C Extraction of Transmission Line Parameters
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Acknowledgements 107
Acknowledgements
I would like to sincerely thank my Professors Dr.-Ing. Reinhold No`
e and Dr.-Ing. Andreas Thiede
for offering me an opportunity to work in their internationally recognized groups in the area of
optical communication and high-frequency electronics. I am grateful for their generous support and
motivating discussions during my dissertation. I sincerely thank both of them for their kindness,
patience and encouragement.
Further, I would like to thank Professor Dr. math Friedhelm Meyer auf der Heide for acting as
supervisor during my work.
I would like to thank Dr.-Ing. David Sandel and Dr.-Ing. Suhas Bhandare from whom I received
enormous help during the entire course and especially in the last phase of my work.
Here, I would like to thank Mr. Bernd Bartsch and Mr. Gerhard Wieseler for their constant
encouragement and technical support.
I gratefully acknowledge the help that I received from all my colleges and friends from the
Optoelectronic and High Frequency Electronic Group at Electrical Engineering department of the
University of Paderborn - Selvan K. Ibrahim, Ariya Hidayat, Sebastian Hoffmann, Abas Ahmad
Fauzi, Dr. Olaf Adamczyk, Vitali Mirvoda, Zheng Gu and Dr.-Ing. Frank W¨
ust.
I would also like to thank my husband and my son Nikola for continuous support, in spite of the
long periods of my physical and mental absence they had to put with. I also would like to thank my
parents for giving constant moral support throughout the entire stay in Germany.
Paderbron, Germany Biljana Milivojevic
April 7, 2005