Dual-band differential filter using broad-band common-mode rejection artificial transmission line

Progress In Electromagnetics Research, Vol. 139, 779–797, 2013 DUAL-BAND DIFFERENTIAL FILTER USING BROADBAND COMMON-MODE REJECTION ARTIFICIAL TRANSMISSION LINE Armando Fern´andez-Prieto1, Jes´us Martel1, Francisco Medina1, *, Francisco Mesa1, Shilong Qian2, Jia-Sheng Hong2, Jordi Naqui3, and Ferran Mart´ın3 1Microwaves Group, Department of Electronics and Electromagnetism and Department of Applied Physics 1 and 2, University of Seville, Av. Reina Mercedes s/n, 41012 Seville, Spain 2Department of Electrical, Electronic and Computer Engineering, Heriot-Watt University, Edinburgh, UK 3GEMMA/CIMITEC, Departament d’Enginyeria Electr´onica, Universitat Autonoma de Barcelona, Spain Abstract—A new balanced dual-band bandpass filter with strong common-mode rejection is presented in this paper. Common-mode rejection is provided by a section of a periodic microstrip differential line that behaves as a low-pass filter under common-mode operation. In contrast, the differential line exhibits very good all-pass behavior under differential mode operation. This structure is combined with a differential dual-band bandpass filter based on embedded resonators. Simulations and experiments confirm that the combined structure has good common-mode rejection within the passbands of the dual-band differential filter. 1. INTRODUCTION Differential signaling plays an essential role in the design of modern high-speed digital circuits because of its comparatively high degree of immunity to noise coming from electromagnetic interference (EMI) and crosstalk. Ideally, differential signal transmission methods meet the tight requirements of EMI compliance and signal integrity but, in practice, any differential signal has some level of common-mode Received 14 April 2013, Accepted 10 May 2013, Scheduled 21 May 2013 * Corresponding author: Francisco Medina ([email protected]). 780 Fern´andez-Prieto et al. noise (mainly caused by amplitude unbalance and time skew). Since this common-mode level can lead to undesired radiation and EMI problems that degrades the performance of high-speed circuitry, a lot of effort has been devoted to the design of differential devices with high common-mode rejection [1–8]. A different approach to achieve good performance is to combine artificial transmission lines providing good common-mode suppression (while keeping the integrity of the differential path) with differential devices (such as bandpass filters) whose common-mode response might not be satisfactory. In the MHz domain, common-mode chokes have traditionally been used to suppress common-mode propagation [9, 10], although such strategy does not work in the GHz domain. In this domain, it was proposed in [11] the use of a LTCC compact narrow band common-mode rejection filter based on coupled lumped inductors. However, the method in [11] makes use of a laborious fabrication process that involves a complicated multilayer structure with via-holes. Alternatively, simpler structures based on etched defected ground structures (DGS) can be used to suppress the common-mode signal. Dumbbell [12] and UH-shaped [13] DGSs have been periodically etched in the ground plane, below a pair of coupled strips signal line, to reject the common-mode noise. Those DGSs can be seen as parallel LC resonators for the common-mode signals and ideal shorts for the differential signals. Since the common-mode current is forced to flow through the ground plane, the etched defect has a significant influence on that mode whereas the differential mode is slightly affected. Nevertheless, the introduction of slots in the ground plane modifies the differential-mode impedance, leading to some level of mismatch losses for the differential signals. Another metamaterialinspired solution has recently been demonstrated in [14]. This proposal makes use of complementary split-ring resonators in the ground plane to yield very strong common-mode rejection over a relatively narrow band with no perturbation of the differential mode. (a) (b) εr Figure 1. Top (a) and bottom (b) metallization patterns corresponding to the common-mode suppression structure analyzed in this paper. Four cells are shown. Black areas correspond to metallized regions. Progress In Electromagnetics Research, Vol. 139, 2013 781 In this paper, the broadband common-mode filtering transmission structure for GHz differential signals introduced in [15] is studied in depth and used to build a compact high-performance balanced dualband filter. Dual-band [16–22] and multiband [23–25] components are becoming more and more important in modern wireless systems. Therefore, the implementation of the differential versions of such filters is of great importance [26]. A sketch of the common-mode rejection structure studied in this paper is depicted in Fig. 1 (top and bottom side views). A pair of coupled microstrip lines is loaded with a periodic distribution of centered conductor patches. The patches are symmetrically series connected to the ground plane by means of narrow (high impedance) strip lines. These lines behave as inductors in the frequency range of interest. Under common-mode operation, the pattern printed in the ground plane in conjunction with the coupled lines of the top side behave as a low-pass filter. Common-mode signal is then expected to be strongly rejected over a wide frequency band above a certain adjustable frequency. Simultaneously, the mismatch of the differential line caused by the presence of the slotted ground plane is minimized by properly tuning the dimensions of lines and patches at both sides of the substrate. Thanks to the broadband behavior of that structure, its combination with a simple balanced dual-band filter (with intrinsic poor common-mode rejection) will allow us to implement a balanced dual-band differential filter with good commonmode performance. The dual-band geometry used in this paper is inspired in the single-ended design in [27]. (a) (b) LS CP L p A A l w w swc w L lc lL wr Top view Bottom view sg 2Cs2Cs LS 1’ 4 2 3 1 3’ 3 2’ 21 4’ 4 ' Figure 2. (a) Top and bottom views of a single unit cell of the proposed artificial transmission line (gray areas are the printed conductors). (b) Equivalent circuit for common-mode operation (AA0 plane is a magnetic wall). The correspondence between physical and circuit model ports is indicated in the drawings. 782 Fern´andez-Prieto et al. 2. ANALYSIS In the double-side transmission line proposed for common-mode suppression in Fig. 1, a number (four in the figure) of identical rectangular patches connected to ground through narrow strips are etched in the ground plane side, below the pair of differential microstrip lines. A detail of the basic unit-cell of this structure is shown in Fig. 2(a). Note that the structure is symmetrical with respect to the AA0plane. In the frequency range of interest, the narrow meandered lines are intended to behave as quasi-lumped inductors. The pair of microstrip lines on the top side of the substrate are capacitively coupled to the patches. 2.1. Circuit Modeling of the Common Mode Under common-mode excitation of the coupled strip lines (even mode using the terminology for single-ended circuits), the AA0plane is a magnetic wall and the current is then forced to flow partially through the LC resonators. This situation can be modeled by means of the equivalent circuit in Fig. 2(b) (common-mode operation). The circuit consists of two series inductances (Ls) and a shunt to the ground LC series branch (Lp,Cp). The capacitance Cptakes into account the capacitive coupling between the top and bottom metalization levels, Lpaccounts for the inductance of the meandered lines, and Lsis the inductance of the top side printed strips. Two additional capacitors, Cs, have been included to account for the electric coupling between closely spaced rectangular patches (the separation gap is sg). If sgis large enough, Cscan be neglected, thus obtaining a simple equivalent low-pass 2-port network. The structure would reject common-mode signals above the resonance frequency of the shunt connected LC- series branch. If the effect of Csis not negligible, we should use a 4-port network description to analyze the influence of the DGS on common-mode operation. Either way, the structure provides strong rejection of the undesired common-mode noise over a relatively wide (in comparison to other reported solutions [12–14]) frequency band. Note that the equivalent circuit in Fig. 2(b) resembles the circuit proposed in [28] for the even mode, but the inductance Lpreplaces here a resonant LC circuit in [28]. This resonant circuit makes the solution in [28] inherently narrower band. Our proposed structure would be useful as a differential transmission system provided the differential (odd) propagating mode remains unaffected by the presence of the DGS pattern. Since the AA0plane behaves as a perfect electric conductor (odd mode), the return path for the odd mode current is only slightly affected by the Progress In Electromagnetics Research, Vol. 139, 2013 783 ground plane modifications. Indeed, current flowing through high impedance meandered lines is negligible because these inductors are short-circuited (physically or virtually) at their two ends. Most of the current flows through the straight coupled lines, with only a small portion flowing over the rectangular patches and through the gaps between them. Therefore, under differential mode operation, the structure will behave as an all-pass continuous transmission line with the characteristic impedance corresponding to the differential mode. In order to verify the above qualitative reasoning, three prototypes have been fabricated on a substrate of thickness t= 0.49 mm and εr= 2.43. The geometrical dimensions (see Fig. 2 for notation) are: w= 1.4 mm, s= 2.4 mm, l= 5.4 mm, wc= 5.2 mm, lc= 5 mm, wL= 0.2 mm, lL= 24.2 mm, wr= 14.6 mm, sg= 0.2 mm. The values of some of these dimensions (w,s,wc, and wr) have been obtained using the quasi-static algorithm in [29]. They have been adjusted to satisfy the 50 Ω odd-mode characteristic impedance condition (100 Ω for the differential mode). This choice minimizes mismatching effects when the differential signal passes from the unpatterned to the patterned ground plane regions. Since that condition can be satisfied using various sets of dimensional parameters, the width, w, of the signal strips has been forced to be the one providing 50 Ω characteristic impedance for the isolated microstrip line printed on the used substrate. Simulated (Agilent ADS Momentum) and measured common-mode insertion losses for three implementations (one, two, and four cascaded cells) of the proposed common-mode filtering section are shown in Fig. 3. The responses predicted by the equivalent circuit models (the circuit models are included in Fig. 3) have been represented for comparison purposes. The values of the circuit components have been extracted for the single unit cell. More specifically, Cphas been roughly approximated as the capacitance of a parallel plate capacitor of plate dimensions l×wwith the dielectric substrate between the plates; Cshas been approximated as the series gap capacitance of the equivalent π-circuit of a microstrip gap discontinuity between two strips of width wseparated a distance sg[30]; the value of Lphas been extracted from the location of the transmission zero close to 2 GHz. Note that the location of this zero can be tuned by controlling the length lLof the meandered line inductor. Finally, Lsis chosen to fit the transmission level in the common-mode passband. The values of the lumped parameters for this case example are found to be: Ls= 4.20 nH, Cp= 0.35 pF, Cs= 0.022 pF, and Lp= 16 nH. Due to the capacitive coupling between the inner conductor patches, the single transmission zero in Fig. 3(a) is split into two or three zeros when two or four unit cells are cascaded. Note that 784 Fern´andez-Prieto et al. (c) 0123456 -50 -40 -30 -20 -10 0 frequency (Ghz) (b) 012345 6 -50 -40 -30 -20 -10 0 frequency (Ghz) || (dB) S21cc 012345 6 -60 -50 -40 -30 -20 -10 0 frequency (Ghz) || (dB) S21cc ||(dB)S21cc (a) Ls1 Ls1 Ls1 Ls1 Ls1 2Ls1 2Ls1 2Ls1 2Ls1 Ls1 Cp2 Cp2 Cp2 Cp2 Cp2 Cp2 Cp2 CsCsCs Cs CsCs CsCs Cs Cs Lp2 Lp2 Lp2 Lp2 Lp2 Lp2 Lp2 Figure 3. Full-wave simulated (solid black lines) and measured (blue dashed lines) common-mode responses for: (a) a single unit cell, (b) two cascaded unit cells, (c) four cascaded unit cells. Equivalent circuits and their predicted behavior (red dotted lines) are included. Csapproximately stands for the coupling to ground capacitors for the first and last patches. In the case of a single unit cell, the equivalent common-mode lumped circuit would strictly behave as a bandstop section rather than a low-pass section. However, the value of Csis very small thus pushing the upper cutoff frequency pole towards a very high frequency value (18.5 GHz in the case treated in this example). Therefore, even taking into account the capacitors connected to ground, we can achieve a wide rejected frequency band. As expected, increasing the number of cells makes higher the rejection level of the common mode, at least in the range of frequencies for which the lumped equivalent circuit is reasonably valid. The insertion losses for the rejected band are above 6 dB for the single unit cell, 14 dB for the two unit-cells configuration, and 20 dB (2–4 GHz range) for the case with four unit cells. A reasonably good agreement can be found between measurements, EM simulations, and circuit simulations Progress In Electromagnetics Research, Vol. 139, 2013 785 012345 6 -60 -48 -36 -24 -12 0 -10 -8 -6 -4 -2 0 | |(dB) Sdd11 ||(dB)Sdd21 frequency (GHz) ||Sdd11 ||Sdd21 Figure 4. Measured (dashed lines) and calculated (solid lines) differential mode insertion (|Sdd21|) and return (|Sdd11|) losses of four cascaded cells of the proposed structure (black curves). The measured (dashed) and calculated (solid) response for a conventional pair of coupled lines (blue curves) has been included for comparison. in Fig. 3. The results provided by the equivalent circuit seem to be less accurate at high frequencies as the number of cells is increased. Thus, although the use of four unit cells increases the rejection level at low frequencies, the high frequency behavior is deteriorated. The results in this paper are compared in Table 1 wit other structures proposed in the literature. It can be seen that our proposal provides the largest fractional bandwidth while in a compact area. Table 1. Comparison of several differential lines providing common mode suppression. Ref. Length Width Area FBW (%) (λe) (λe) (λ2 e)−20 dB [12] 0.76 0.47 0.36 50 [13] 0.44 0.44 0.19 — [14] 0.64 0.13 0.08 41 [28] 0.26 0.16 0.04 32 This 0.32 0.22 0.07 67 In Fig. 4 we show the measured and simulated differential mode transmission for the four cells structure in Fig. 3 and for a simple pair of differential microstrip lines without ground plane etching. From 786 Fern´andez-Prieto et al. the measured data in Fig. 4 we can conclude that the presence of the patches and meandered lines in the ground plane side of the substrate does not meaningfully affect the differential mode operation, as expected and desired. Note that transmission degrades at relatively high frequencies in both cases (patterned and unpatterned ground planes). This is due to differential-to-common mode conversion associated to unavoidable unbalanced loading in our measurement setup. This degradation is not observed in the full-wave simulations since in such case symmetries are perfect. 2.2. Bloch-Floquet Analysis The common-mode rejection behavior can be explained in terms of the dispersion equation for the Bloch waves supported by a periodic structure consisting of the cascade connection of an infinite number of identical unit cells (as the one depicted in Fig. 2(a)). This Bloch- Floquet analysis has been carried out, for instance, in [14, 28] to explain the stopband common-mode operation of the proposed structures in terms of the forbidden band of the periodic structures. In order to apply a similar analysis to the structure proposed in Fig. 1, we first carry out a simplified study where it is neglected the coupling capacitor between ground plane patches. If Cs= 0, the circuit in Fig. 2(b) becomes a simple 2-port network whose dispersion relation can be obtained from the ABCD transmission matrix of the unit cell [31, Chap. 8] as cosh(γl) = A. (1) In the above equation, γis the complex propagation constant of the Bloch wave (γ=α+jβ), lis the length of the unit cell, and the element Aof the transmission matrix is given by A=1−ω2Cp(Lp+Ls) 1−ω2LpCp (2) where ωis the angular frequency. For the previously extracted values of the circuit parameters, (1) and (2) yield β= 0 and α6= 0 (no propagation, only attenuation) from 2.1 GHz onwards; namely, there is not an alternating sequence of allowed/forbidden bands as in [14, 28]. The structure behaves as a low-pass filter rather than as a bandstop filter. However, it should be mentioned that the distributed effects not included in this simplified circuit model will modify this prediction at frequencies above the range of interest of this paper. Nevertheless, the low-pass filtering behavior here obtained qualitatively explains why our proposed structure provides good common-mode rejection over a wide frequency band. Progress In Electromagnetics Research, Vol. 139, 2013 787 Unfortunately, to ignore the nonvanishing value of Csmight be not very realistic. Actually, closely spaced patches in the bottom side of the substrate are desirable since they would provide a better differential mode performance. If the interaction between adjacent patches is accounted for by a non-negligible edge coupling capacitance (Cs6= 0), the resulting equivalent-circuit unit cell should be treated as a 4-port network, as suggested in [14, 32] for other structures. Starting from a 4-port ABCD matrix formulation, following the guidelines in [14, 32], the dispersion equation can be written as follows: det{[A]−cosh(γl)[I]}= 0 (3) where [I] is the unit matrix and [A] the 2 ×2 matrix that relates the voltages of the left (input) side of the 4-port network in Fig. 2(b) [V1 and V3] to the voltages of the right (output) side [V2and V4] provided the right side ports are open circuited; namely, ·V1 V3¸=·A11 A12 A21 A21¸·V2 V4¸for I2=I4= 0.(4) For the 4-port network in Fig. 2(b), the elements of the matrix [A] are readily found to be A11 = 1 −ω2LsCp(5) A12 =ω2LsCp(6) A21 =−Cp 2Cs (7) A22 = 1 + Cp 2Cs −1 ω2LpCs .(8) The solution of Eq. (3) leads to the two following modal solutions for γ: cosh(γl) = 1 2³A11 +A22 ±p(A11 −A22)2+ 4A12A21´.(9) It should be noted that the two solutions come from the ±choice in (9) and that, if γis a solution of (9), −γwill also be another solution. In Fig. 5 we have plotted the behavior of the positive real and imaginary parts of the two modal propagation constants. In the lowfrequency region up to ∼1.8 GHz, one mode is propagative (blue-color solution with α= 0, β6= 0) and the other is evanescent (red-color solution with α6= 0, β= 0). Above 3.0 GHz, the two modes are purely evanescent (α6= 0, β= 0) and then they do not carry active power. In the middle interval (1.8–3.0 GHz), the 4-port equivalent circuit predicts the existence of complex modes (α6= 0, β6= 0). Since complex modes are excited by a single source in pairs and each pair 794 Fern´andez-Prieto et al. 5. CONCLUSIONS In this work, an approach for the design of differential lines with common-mode suppression in the gigahertz frequency range has been presented. The differential transmission lines have been implemented using double-side microstrip technology. 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