Tunable Multiphase Oscillator Using Diamond Transistors with Voltage Controlled Condition of Oscillation for Amplitude Stabilization

ELEKTRONIKA IR ELEKTROTECHNIKA,ISSN 1392-1215,VOL.20,NO.1,2014 1Abstract—The main aim of this paper is to present simple and low-cost solution of oscillator utilizing very simple method for automatic amplitude stabilization using adjustable current gain control. It provides satisfying total harmonic distortion results and adjusting of oscillation frequency over one decade. Experimental results confirmed workability and gave opportunity to consider and evaluate practically achievable performances of this oscillator type. Index Terms—Electronic control, automatic gain control circuit, amplitude stabilization, quadrature and multiphase oscillator, current amplifier, negative resistor. I. INTRODUCTION A sinusoidal oscillator is very important part of communication systems and applications. There are many active elements suitable for construction of oscillators [1]. However, many hitherto published types are not studied in the point-of-view of output signals with low total harmonic distortion (THD) and stable amplitudes. There are many works where stabilization of output amplitudes was not discussed or explained and THD is really terrible [2]. There are some ways how to implement circuit for amplitude stabilization or electronic control of oscillation frequency. The easiest way is to use adjustable resistor (replaced by FET (Field-effect transistor) transistor [3], [4]). It is useful in cases of grounded passive elements in the oscillator circuit. Using of FET as a replacement of floating resistor(s) is quite complicated [4]. An opto-coupler should be a better choice if there are floating elements to be controlled as shown in [5]–[7]. Diode limiters are sufficient in some oscillator structures that are based on highly selective filtering band-pass structures (high quality factor) [8]. Utilization of parameters of active elements Manuscript received January 28, 2013; accepted June 14, 2013. This research work is funded by projects SIX CZ.1.05/2.1.00/03.0072, EU ECOP EE.2.3.20.0094, CZ.1.07/2.2.00/28.0062, and by Czech Science Foundation project 102/09/1681, Czech Republic. The support of the project CZ.1.07/2.3.00/20.0007 WICOMT, financed from the operational program Education for competitiveness, is gratefully acknowledged. The described research was performed in laboratories supported by the SIX project; the registration number CZ.1.05/2.1.00/03.0072, the operational program Research and Development for Innovation. (transconductance, intrinsic resistance, gain) is also suitable for control of oscillation condition. Our contribution deals with very simple utilization of current gain of current-mode multiplier in oscillator design which is driven by output voltage. It ensures low THD and quite stable output levels in quite wide range of oscillation frequency adjusting. II. OSCILLATOR STRUCTURE Proposed oscillator uses integrators in the loop complemented by negative resistance which was realized by current-mode multiplier. Similar principle was used with transconductors (OTA) only, in [9], [10], for example. However, solution from [9], [10] is only quadrature (inverting output is not available), uses different (noninertial) AGC (automatic gain control) and principle of CO (condition of oscillation) control (control is performed by nonlinearity of positive resistor). Block diagram that explains our modification is shown in Fig. 1. Current-mode multiplier connected as controllable current amplifier (CA) allows electronic control of current gain in order to adjust resistance to negative value [11]. This is the most important part of the AGC circuit. Principle of operation is briefly given in Fig. 2. Symbol RXrepresents intrinsic resistance of current input of current amplifier. In our case, the current gain Bis given by DC control voltage VG(for EL2082 type of multiplier was used, BVG[11]). Input resistance is given by following equation *4 4, 1 R RB  (1) which leads to */ 4 4 495, 1 1 X G R R R RB V       (2) in case when real behavior is considered (RX95 in case of EL2082 [11]). Value of R4*is negative for VG> 2. Overall circuit diagram of proposed oscillator is shown in Fig. 3. It also contains values of parameters (passive elements) and detailed principle of AGC circuit employing Tunable Multiphase Oscillator Using Diamond Transistors with Voltage Controlled Condition of Oscillation for Amplitude Stabilization J. Jerabek1, R. Sotner2, K. Vrba1 1Department of Telecommunications, Brno University of Technology, Technicka 12, 616 00 Brno, Czech Republic 2Department of Radio Electronics, Brno University of Technology, Technicka 12, 616 00 Brno, Czech Republic [email protected]br.cz http://dx.doi.org/10.5755/j01.eee.20.1.6166 45 ELEKTRONIKA IR ELEKTROTECHNIKA,ISSN 1392-1215,VOL.20,NO.1,2014 one bipolar transistor which produces DC voltage for control of current gain B(negative resistance value). Increase of output level causes higher base-emitter voltage of BJT (bipolar-junction transistor) and therefore decrease of collector voltage and B of CA (R4* value increases). Decreasing output level of the oscillator causes opposite reaction of this inertial AGC loop. This method uses nonlinear dependence of collector voltage on base-emitter voltage of BJT. Fig. 1. Block diagram of proposed oscillator structure. Fig. 2. Structure of grounded negative resistance made from current amplifier. Fig. 3. Complete circuit diagram of the oscillator. Two integrators are constructed from two diamond transistors [12] and time constants are controllable (simultaneously) by two grounded resistances, R1and R2, in accordance to block diagram from Fig. 1. Very simple oscillators using DT-s (diamond transistors) were investigated in [13] but qualitative tunable features (low THD and unchangeable output levels) were not the main aim of contribution. All voltages in high-impedance nodes (at C1, C2) are available through voltage buffers (part of OPA860 package [14]) and therefore are available at low-impedance outputs without any further circuit complexity. Inverting voltage buffer (created by AD8138 [15]) is connected between integrators and provides also availability of lowimpedance output of signal with 180 degree phase shift. Routine analysis of characteristic equation provides the following results   * 1 3 4 2*1 2 1 2 234 10, C R R s s R R C C C R R     (3) where R4*can be replaced by (1) in ideal case 4 1 3 2 41 2 1 2 2 3 1 10. 1 R C R B s s RR R C C C R B             (4) Final form of characteristic equation is   1 3 3 4 2 2 3 4 1 2 1 2 110, C R B R s s C R R R R C C          (5) where we can find oscillation condition and oscillation frequency very easily as: 4 3 1 , RB R   (6) 01 2 1 2 1. R R C C  (7) Relations between generated voltages achieve -90 and 180 degrees as we can determine from: 2 2 1 2 1 1 , OUT OUT R C V j V R C   (8)   0 12 2 2 2 2 2 1 1 1 1 j arctg j OUT OUT VR C R C e e V R C R C      (9) and from 1 3 3. j OUT OUT OUT V V e V     (10) Relative sensitivities of oscillation frequency (0,f0) on all influencing parameters are −0.5, as obvious from (7). Simultaneous control of R1and R2(for example by digital potentiometers) allows linear control of f0. III. EXPERIMENTAL VERIFICATION We utilized commercially available active elements for experimental laboratory tests of designed oscillator. It allows fast, appropriate, efficient and low-cost analysis of real behavior. A. Discussion of Main Parasitic Influences The most important (for f0accuracy) are RE13 [14] (intrinsic resistance of emitter port of the DT), output (collector) resistance of DT (it is the lowest value in high impedance nodes RC1,2 54 k) and capacitances in high impedance nodes as total sum of collector capacitance 46 ELEKTRONIKA IR ELEKTROTECHNIKA,ISSN 1392-1215,VOL.20,NO.1,2014 (DT2), base capacitance (DT1) and output capacitance of current-mode multiplier [11] (CB1 = CC2 2 pF; Cinp_buffer2 2 pF; CZ_multiplier 5 pF) in node C2and total sum of collector capacitance (DT1) and inverter input capacitance [15] (CC1 2 pF; Cinp_buffer1 2 pF; Cinp_inverter 1 pF) in node C1. B. Design of Oscillator Design parameters are: R1=R2=R<119; 9783 >  (parasitic values of REof DT are taken into account [14]), C1=C2=C= 470 pF, R3=R4= 1 k. Range of Rvalue adjusting was calculated from (7). Adjustable trimmers allow careful setting of oscillation condition. C. Measured Results Comparison of measured and ideal dependence of oscillation on simultaneous change of both R1=R2=Ris shown in Fig. 4. Dependences of achieved output levels on frequency are in Fig. 5. 1.0E+04 1.0E+05 1.0E+06 1.0E+07 100 1000 10000 R1,2 [W ] f0[Hz] measured ideal VCC = ± 5 V Fig. 4. Dependence of oscillation frequency on resistance value of tandem potentiometer. 2 2,5 3 3,5 0,0E+00 5,0E+05 1,0E+06 1,5E+06 2,0E+06 2,5E+06 3,0E+06 f0 [Hz] VOUT1,2[Vp-p] VCC = ± 5 V OUT1 OUT2 Fig. 5. Output levels vs. oscillation frequency. Figure 6 shows progression of total harmonic distortion (THD), which is sufficiently low (around 0.5 %) almost in whole range of f0adjusting. Because OUT3is only inversion of OUT1analyses for OUT1and OUT2are sufficient (output levels and parameters are the same). We achieved ideal range of an f0adjusting from 35 kHz to 2.859 MHz. Measured results showed range from 40 kHz to 2.510 MHz. Equation (6) allows calculation of sufficient Bfor start of oscillations. In our case it is B= 2. Response of automatic gain control circuit (controlling voltage VGrespectively) on oscillation frequency and changes of oscillation condition in the loop (Barkhausen criterion [16]) are shown in Fig. 7. 0 0.5 1 1.5 2 2.5 3 0.0E+00 5.0E+05 1.0E+06 1.5E+06 2.0E+06 2.5E+06 3.0E+06 f0 [Hz] THD [%] VCC = ± 5 V OUT2 OUT1 Fig. 6. THD vs. oscillation frequency. 1.0 1.2 1.4 1.6 1.8 2.0 2.2 1.0E+04 1.0E+05 1.0E+06 1.0E+07 VG [V] f0[Hz] VCC = ± 5 V Fig. 7. Response of AGC (VGcontrol of current-mode multiplier) on tuning performance. Fig. 8. Transient responses (CH1OUT1, CH2= OUT2, CH3= OUT3) for R= 119 and VG= 1.55 V (f0= 2.514 MHz). (a) 47 ELEKTRONIKA IR ELEKTROTECHNIKA,ISSN 1392-1215,VOL.20,NO.1,2014 (b) Fig. 9. Examples of FFT spectrum of a) OUT1and b) OUT2; for R= 119  and VG= 1.55 V (f0= 2.514 MHz). Higher f0frequencies mean higher gain in the open-loop circuit and therefore AGC causes decreasing of sufficient B to the lower values. Transient response of available output signals and FFT spectrum in both high-impedance nodes (C1,C2after buffering) are shown in Fig. 8 and Fig. 9. Suppression of higher harmonic components is higher than 48 dBc. IV. CONCLUSIONS In this contribution, adjustable oscillator was presented. Oscillation condition is controlled by simple AGC employing current gain control and using nonlinearities of bipolar transistor and simple diode doubler for detection of control magnitude from output voltage. In combination with suitable oscillator structure, this approach allows to achieve wideband and linear oscillation frequency control. In our case, we achieved range of adjusting 1:60. Adjusting in range where THD is low (around 0.5%) achieves 1:25 approximately. REFERENCES [1] D. Biolek, R. Senani, V. Biolkova, Z. 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