ECCCII-Based Current-Mode Universal Filter with Orthogonal Control of w_o and Q

RADIOENGINEERING, VOL. 23, NO. 2, JUNE 2014 687 ECCCII-Based Current-Mode Universal Filter with Orthogonal Control of  o and Q Montree KUMNGERN 1, Fabian KHATEB 2,3, Pattarapong PHASUKKIT 1, Supan TUNGJITKUSOLMUN 1, Somyot JUNNAPIYA 1 1 Faculty of Engineering, King Mongkut’s Institute of Technology Ladkrabang, Bangkok 1520 Thailand 2 Dept. of Microelectronics, Brno University of Technology, Technická 10, Brno, Czech Republic 3 Czech Technical University in Prague, Faculty of Biomedical Engineering, Nám. Sítná 3105, Kladno, Czech Republic [email protected], [email protected], [email protected], [email protected], [email protected] Abstract. This paper presents a new current-mode current-controlled four-input five-output universal filter employing one current-controlled current conveyor (CCCII), one electronically tunable CCCII and two grounded capacitors. The proposed configuration provides lowpass, bandpass, highpass, bandstop and allpass current responses that taken from the high-output impedance terminals, which enable easy cascadability of the current-mode operation. The filter also offers both orthogonal and electronic controls of the natural frequency and the quality factor through adjusting the bias current of the CCCIIs. For realizing all the filter responses, the proposed filter does not require passive component-matching condition and both active and passive sensitivities are low. In addition, a new current-mode current-controlled single-input five-output universal filter can be achieved by using an additional multiple-output minus-type CCCII. The proposed filter is simulated using PSPICE simulations to confirm the theoretical analysis. Keywords Universal filter, current-mode circuit, electronically tunable current conveyor, analog filter. 1. Introduction In the present, current-mode signal processing circuits based on second generation current-conveyors (CCIIs) have received considerable attention owing to the fact that their bandwidth, linearity and dynamic range performances are better than op-amp-based voltage-mode signal processing circuits [1], [2]. Especially, current-mode circuit provides simple summation/subtraction signals of currents in node, which results to simple circuitry. Many currentmode universal filters using current conveyors as active elements have been proposed [3]-[43]. A number of CCIIbased current-mode universal filters have been presented [3]-[9]. In [3]-[6] single-input multiple-output (SIMO) current-mode universal filters using CCIIs have been reported. Generally, SIMO filter can simultaneously realize three basic filter functions, i.e. lowpass (LP), bandpass (BP) and highpass (HP). However, for the realizations of allpass (AP) and bandstop (BS) functions, some of SIMO filters require passive or signal-matching condition. For more convenience and versatility, the multiple-input multiple-output (MIMO) universal filters could be used. The employment of the MIMO configuration may lead to a reduction of number of active elements for circuit realization. This type of filter, in comparison with the SIMO filter, provides a variety of circuit characteristic with different input and output currents, and usually does not require any parameter matching conditions. Moreover, to realize all the standard biquadratic filter functions, the configuration with multiple inputs seems to be more suitable than the single input configuration [11]. In [7]-[9] CCIIs-based MIMO current-mode universal filter have been presented. Most of the MIMO universal filters based on CCIIs suffer from lack of electronic tuning. By using the second-generation current-controlled current conveyor (CCCII) introduced by Fabre et al. [10], current conveyor applications can be extended to the domain of electronically tunable functions. Many current-mode CCCII-based universal filters have been proposed in the technical literatures [11]- [42]. In [11]-[16] multiple-input universal filters were proposed while single-input universal filters were proposed in [17]-[41]. However, some of these filters do not benefit from orthogonal control of the natural frequency  o and the quality factor Q. In this paper, we propose a new electronically tunable current-mode four-input five-output universal filter using one CCCII, one electronically tunable CCCII and two grounded capacitors [42], which is advantageous in view of integrated circuit implementation. The proposed circuits can simultaneously realize LP, BP, HP, BS and AP current responses at a high impedance output terminal permitting easy cascadability of the current-mode operation. The circuit parameters  o and Q can be tuned separately and electronically by adjusting the bias current of CCCII. Moreover, high Q-value filters can be obtained. For the realization of all the filter responses no component-matching conditions are required. Both active and passive sensitivities are low. In addition, a new current-mode single-input 688 M. KUMNGERN, F. KHATEB, P. PHASUKKIT, S. TUNGJITKUSOLMUN, S. JUNNAPIYA, ECCCII-BASED CURRENT-MODE... five-output universal filter can be also achieved by adding a multiple-output minus-type CCCII into the proposed filter. This new circuit has low-input and high-output impedance levels. The comparison between the proposed circuits and some previously CCCII-based filters is summarized in Tab. 1. 2. Proposed Circuit The well-known schematic for CCCII, implemented with bipolar technology is shown in Fig. 1 [10]. According to Fig. 1, the CCCII has a unity voltage gain between terminal y and x and a unity current gain between terminal x and z. The terminals y and z possess high impedance level and the x terminal has the Rx, which can be given by 2 T x o V RI  (1) The Rx is an inner resistance of a translinear mixed loop (Q1 to Q4) with grounded resistor equivalent controlled by bias current Io, where VT is the thermal voltage ( 25 mV). The translinear current conveyor with controlled current gain can be obtained by modifying the original circuit of the CCCII in Fig. 1 and adding additional current mirror with adjustable gain as shown in Fig. 2 [44] to obtain the required current gain at z terminal. Also, the multiple-output translinear current conveyor can be obtained by adding additional current mirrors and cross-coupled current mirrors to obtain the required plus and minus type outputs, respectively [45]. Fig. 1. Schematic implementation for CCCII. Circuits Type of filter Number of active elements Number of capacitors Allgrounded capacitor Offer five standard filters Orthogonal control of  o and Q No need matchingcondition Low active & passive sensitivity High output impedance Proposed filters Fig. 3(a) Fig. 3(b) MIMO SIMO 2 CCCIIs 3 CCCIIs 2 2 Yes Yes Yes Yes Yes Yes Yes Yes Yes Yes Yes Yes Ref. [11] in 2006 MISO 5 CCCIIs 2 Yes Yes Yes No Yes Yes Ref. [12] in 2007 MIMO 4 CCCIIs 2 Yes Yes Yes No Yes Yes Ref.[14] in 2007 MIMO 2 CCCIIs 2 Yes Yes No Yes Yes Yes Ref. [15] in 2010 MIMO 2 CCCIIs 2 Yes Yes Yes No Yes Yes Ref. [16] in 2008 (fig. 3) MISO 2 CCCII 2 Yes Yes No No Yes Yes Ref. [19] in 1997 SIMO 3 CCCIIs 2 Yes No No No Yes Yes Ref. [20] in 1998 SIMO 6 CCCIIs 2 Yes No No No Yes Yes Ref. [21] in 1998 SIMO 3 CCCIIs 2 Yes No No Yes Yes No Ref. [23] in 2000 SIMO 3 CCCIIs 2 Yes No Yes Yes Yes No Ref. [26] in 2004 SIMO 4 CCCIIs 2 Yes No No Yes Yes Yes Ref. [27] in 2004 SIMO 2 CCCIIs 2 No No No Yes Yes No Ref. [28] in 2005 SIMO 3 CCCIIs 2 Yes No No Yes Yes Yes Ref. [29] in 2006 SIMO 2 CCCIIs 2 No No No Yes Yes No Ref. [30] in 2006 SIMO 4 CCCIIs 2 No No No Yes Yes Yes Ref. [31] in 2007 SIMO 3 CCCIIs 2 Yes No No Yes Yes Yes Ref. [33] in 2008 SIMO 4 CCCIIs 2 Yes Yes Yes Yes Yes Yes Ref. [34] in 2009 SIMO 4 CCCIIs 2 Yes No No No Yes Yes Ref. [35] in 2009 SIMO 5 CCCIIs 2 Yes No Yes Yes Yes Yes Ref. [37] in 2009 SIMO 3 CCCIIs 2 Yes Yes No Yes Yes Yes Ref. [38] in 2010 SIMO 3 UCC 2 Yes Yes No Yes Yes Yes Ref. [39] in 2010 SIMO 3 CCCII 2 Yes Yes No Yes Yes Yes Ref. [40] in 2010 SIMO 3 CCCIIs 2 Yes Yes Yes Yes Yes Yes Ref. [43] in 2004 SIMO 1 CCIII 2 No No No Yes Yes No Note: MISO is multiple-input and single-output. Tab. 1. Comparison of proposed filters with those of previous circuits. RADIOENGINEERING, VOL. 23, NO. 2, JUNE 2014 689 (a) (b) Fig. 2. Current mirrors with adjustable current gain: (a) positive-type, (b) negative-type. Fig. 3(a) and (b) shows respectively the schematic and the symbol of the electronically tunable CCCII [42]. It has a unity voltage gain between terminals y and x and tunable k current gain between terminals x and z. The latter property makes it different from the current conveyor and hence the name electronically tunable CCCII (ECCCII). A few electronically tunable current conveyors were already described in the literature [46]-[48]. However, these devices are not suitable for electronic-control of  o and high Q-value biquadratic filter. The schematic of ECCCII in Fig. 3 is characterized by the relationship: 000 10 010 00 y y xx x z z zk IV VRI IV Ik                   (2) The current gain k of the ECCCII can be expressed as [42] a b I kI  (3) It is evident that the signal is amplified by the factor k and this factor can be varied linearly and controlled by adjusting the Ia/Ib. Fig. 4 shows the block diagram of the proposed universal filter. It comprises of two integrators and one amplifier. Based on Fig. 4 the proposed current-mode universal filter is shown in Fig. 5. From Fig. 5(a), CCCII1 and C1 operate as the first integrator (1/s  1) and current amplifier (k) while CCCII2 and C2 operate as the second integrator (1/s  2). The filter in Fig. 5(a) is consisted of one CCCII, one ECCCII and two grounded capacitors and hence the name ECCCII-based universal filter. The use of grounded capacitor makes the proposed filter ideal for integration point of view [49], [50]. Using (2) and nodal analysis, the transfer functions of the proposed biquadratic filter in Fig. 5(a) can be expressed as 2 1212 221 1 -1 LP in xx x I IsRRCC sRCk  , (4) 221 2 1211 221 -1 x BP in xx x sR C k I IsR R CC sR Ck  , (5) (a) Vx Vy Iz Ix Iy CCCII x yz+ kzVz Izk Vz Vzk kz+ Izk Vzk zIz (b) Fig. 3. ECCCII; (a) bipolar implementation; (b) circuit symbol. 690 M. KUMNGERN, F. KHATEB, P. PHASUKKIT, S. TUNGJITKUSOLMUN, S. JUNNAPIYA, ECCCII-BASED CURRENT-MODE... Fig. 4. Block diagram of proposed universal filter. 2 1212 2 1212 221 1 xx HP in xx x sR R CC I IsR R CC sR Ck   , (6) 2 1212 2 1212 221 1 1 BS x x in xx x IsRRCC IsR R CC sR Ck   , (7) 2 1212 221 2 1212 221 -1 1 xx x AP in xx x sR R CC sR Ck I IsR R CC sR Ck   . (8) Therefore, five filter functions can be achieved. It should be mentioned that there is no component-matching condition for realizing all the standard type biquadratic filtering functions. The natural frequency  o and the quality factor Q are given by 1212 1 o xx R RCC  , (9) 11 122 1x x R C QkRC  (10) (a) (b) Fig. 5. Proposed current-mode universal filters: (a) MIMO-type filter, (b) realization for SIMO-type filter. RADIOENGINEERING, VOL. 23, NO. 2, JUNE 2014 691 with 1 1 1 a b I kI  (11) where Ia1 and Ib1 are the bias currents of CCCII1. If we set Rx1 = Rx2, C1 = C2 and substituting (11) into (10), equation (10) becomes 1 1 b a I QI . (12) From (1) and (9), the parameter  o for all filter responses can be electronically tuned by varying Io1 and/or Io2 without affecting the parameter Q. For the Q-value, it can be controlled linearly and separately by adjusting the ratios of the bias currents Ib1 and Ia1, where the high-Q biquad can be realized when the appropriate current gain is chosen. Moreover, the Q-value is also temperature independent. This means that the proposed circuit can work as a current-tunable filter, and its parameters  o and Q can be independently tuned over a wide range. Note that the requirement of the four input currents for realizing all the filter responses is not a major disadvantage. In practice, the multiple input currents can easily be obtained by using an additional multiple-output CCCII with grounded y-terminal. From Fig. 1, an input current Iin is then injected to the terminal x, while the four input currents can be taken from terminals z-. Fig. 5(b) shows the complete realization of universal filter which gives a transfer function of the form given by (4) to (8). From Fig. 5(b), it can be seen that the proposed universal filter employed only three CCCIIs and two grounded capacitors. Moreover, the input signal Iin is connected to the low input impedance terminal of the CCCII. Therefore, the second proposed circuit provides the advantage of having low-input and high-output impedance. 3. Non-ideal Effects To consider the non-ideal effect of a CCCII by taking the non-idealities of the CCCIIs into account, the relationship of the terminal voltages and currents can be rewritten as 000 0 00 00 Y Y jx X X j Z Z kj j Zk IV R VI IV k I                       (13) where  j = 1 -  vj and  vj(  vj « 1) is the voltage tracking error from Vy terminal to Vx terminal of the j-th CCCII,  j = 1 -  ij and  ij(  ij « 1) is the output current tracking error of the j-th CCCII and  kj = 1 -  ij and  ij(  ij « 1) is the output current tracking error of the j-th CCCII. With regards to the effect of parasitic parameters, it can be evaluated using the non-ideal CCCII symbol as Fig. 6. Simplified equivalent circuit of the non-ideal CCCII. X o ω x S Q x S Rx1 -0.5 0.5 Rx2 -0.5 -0.5 1 C  -0.5 0.5 2 C  -0.5 -0.5 1 0.5 -0.5 2 0.5 0.5 1 0.5 0.5 2 0.5 0.5 k1 0.0 -1.0 k1 0.0 -1.0 Tab. 2. Sensitivities of circuit components. shown in Fig. 6. Typically, Ry, Rz and Rzk possess high value parasitic resistance while Cy, Cz and Czk possess low parasitic capacitance. Using (13) and taking into account the non-ideal CCCII characteristics as shown in Fig. 6, the modified demonstrate D(s) of (4)-(8) can be expressed as 2 111 1212 12 2 () k ss Ds k nn n     (14) where 1 11 1 1 x s nRC s         , (15) 2 22 2 1 x s nRC s         , (16) 1 11 1 n RC    , 2 22 1 n R C    , 11 1 112-ykz z CCC C C    , 22 2 1yz CCC C    , 11112- // // nykz z R RR R, 221 // nyz R RR   . Form (14)-(16), the effect of the parasitic parameters of CCCII are dependent on two parasitic poles. The frequency of operation should be larger than ω1 and ω2 for close to ideal operation at high frequency. Using PSPICE simulations, the parasitic elements of the CCCII for Io = 32 µA can be calculated as Cy1 = Cy2 = 2.2 pF, Cz1+ = Cz2-  1.67 pF, Ck1z1- = 1.1 pF, Ry1 = Ry2 = 358 k, Rz2- = 350 k, Rz1+ = 366 k, and Rk1z1- = 151 k For example, if the proposed filter was designed for fo = 1 MHz by choosing C1 = C2 = 0.4 nF and Rx1 = Rx2  394 , then the parasitic pole f1 will be located at approximately 692 M. KUMNGERN, F. KHATEB, P. PHASUKKIT, S. TUNGJITKUSOLMUN, S. JUNNAPIYA, ECCCII-BASED CURRENT-MODE... 7.5 kHz and f2 will be located at approximately 2.1 kHz. Thus, the ideal operation of the filter will be valid for frequencies higher than approximately 10f1 [9], [10]. Assume that ω1 and ω2 » 1, the parameters  o and Q can be rewritten by 1212 1212 o xx R RCC    , (17) 11 212 11 22 1 1x kx RC QkRC     . (18) From (17) and (18), the tracking errors slightly change the natural frequency and the quality factor. However, the parameters  o and Q can still be orthogonally controllable. The incremental sensitivities of the parameters  o and Q are calculated in Tab. 2. It can be indicated that the sensitivities of the proposed filter are low. 4. Simulation Results In order to verify the characteristics of the proposed filter in Fig. 5, PSPICE simulators were carried out. The CCCIIs were performed with the transistor model of HFA3046 as listed in Tab. 3 and the DC supply voltage VCC = -VEE = 3 V. When the ECCCII in Fig. 3 was simulated by PSPICE simulators, the parameters  = 0.998,  + = 0.995 and  - = 1.065 were obtained and other simulated results were presented in Tab. 4. The frequency response of the ECCCII with different bias current Ia of 50, 250, 500, 700 to 1000 μA was investigated and shown in Fig. 7. As an example design, C1 = C2 = 0.4 nF was given. Fig. 8 shows the simulated frequency responses of the filter with Io1 = Io2 = 32 A and Ia1 = Ib1 = 50 A. This setting was designed to obtain the LP, BP, HP and BS filter responses with fo  1.02 MHz and Q = 1. Fig. 9 shows the simulated frequency responses of the gain and phase characteristics of the AP filter at fo  1 MHz. It was clear from both figures that the proposed filter performs five standard biquadratic filtering functions well. In Figs. 8 and 9, the pole frequency of 1 MHz was obtained. The pole frequency was 1 MHz instead of 1.02 MHz owing to the effect of non-ideal CCCIIs. Fig. 10 shows the simulated BP filter response when the DC bias currents Io (i.e. Io = Io1 = Io2) were simultaneously adjusted for the values 15, 25, 40 and 60 A, respectively, while keeping Ia1 = Ib1 = 50 A for Q = 1. This result was confirmed by (9). To demonstrate the current gain of current conveyor tuning of Q, the bias currents were set to be constant at Io1 = Io2 = 32 A and Ia1 = 50 A. Fig. 11 shows the corresponding current characteristics of the BP filter when Ib1 was varied. It is obvious that high values of the Q can be easily obtained from high ratios of current gain as was confirmed by (12). From Figs. 8 to 11 it is evident that the simulation results agree quite well with the theoretical analysis. Therefore, the proposed filter easily obtains separate electronic control of parameter  o and Q and high-Q value filter by using ECCCII. Recently, a new active device, so-called double current controlled current feedback amplifier (DCC-CFA) was proposed [51], [52]. This device provides the current gain that can be used to realize the filter with separate control of parameter  o and Q and highQ value. However, this DCC-CFA was realized using bipolar and CMOS technologies and thus BiCMOS technology was needed for integrated circuit implementation which is difficult as a fabrication process. .model NUHFARRY NPN (IS=1.840E-16 XTI=3.000E+00 EG=1.110E+00 VAF=7.200E+01 VAR= 4.500E+00 BF=1.036E+02 ISE=1.686E-19 NE=1.400E+00 IKF= 5.400E-02 XTB=0.000E+00 BR= 1.000E+01 ISC=1.605E-14 NC=1.800E+00 IKR=5.400E-02 RC=1.140E+01 CJC=3.980E-13 MJC= 2.400E-01 VJC=9.700E-01 FC=5.000E-01 CJE=2.400E-13 MJE=5.100E-01 VJE=8.690E-01 TR=4.000E-09 TF=10.51E-12 ITF=3.500E-02 XTF=2.300E+00 VTF=3.500E+00 PTF=0.000E+00 XCJC=9.000E-01 CJS=1.150E-13 VJS=7.500E-01 MJS=0.000E+00 RE=1.848E+00 RB=5.007E+01 RBM=1.974E+00 KF=0.000E+00 AF=1.000E+00) .model PUHFARRY PNP (IS=1.027E-16 XTI=3.000E+00 EG=1.110E+00 VAF=3.000E+01 VAR=4.500E+00 BF=5.228E+01 ISE=9.398E-20 NE=1.400E+0 IKF=5.412E-02 XTB=0.000E+00 BR= 7.000E+00 ISC=1.027E-14 NC=1.800E+00 IKR=5.412E-02 RC=3.420E+01 CJC=4.951E-13 MJC=3.000E-01 VJC=1.230E+00 FC=5.000E-01 CJE=2.927E-13 MJE=5.700E-01 VJE=8.800E-01 TR=4.000E-09 TF=20.05E-12 ITF=2.001E-02 XTF=1.534E+00 VTF= 1.800E+00 PTF=0.000E+00 XCJC=9.000E-01 CJS=1.150E-13 VJS=7.500E-01 MJS=0.000E+00 RE=1.848E+00 RB=3.271E+01 RBM=9.902E-01 KF=0.000E+00 AF=1.000E+00) Tab. 3. Parameter of HFA3046 transistor array. Parameters Value Power supply  3 V DC voltage range -2 to 2 V Voltage error range <7 mV DC current range -25 to 25 mA Bandwidth (-3dB) @ [Io= 50 µA, Ia= 50 µA, Ib= 50 µA] Voltage follower (Vx/Vy) Current follower (Iz/Ix) Current follower (Izk/Ix) 370 MHz 177 MHz 112 MHz Rx, Lx @ [Io =50 µA] 253 , 0.73 µH Ry, Cy 358 k, 2.2 pF Rz+, Cz+ Rzk, Czk @ [Ia=Ib=50 µA] Rzk, Czk @ [Ia=1000 µA, Ib=50 µA] Rzk, Czk @ [Ia=50 µA, Ib=1000 µA] 366 k, 1.67 pF 150 k, 1.19 pF 13.59 k, 0.39 pF 3.6 M, 0.23 pF Tab. 4. Simulated specifications of CCCII used. In order to test the time-domain of the proposed filter, a LP filter at 1 MHz of cut-off frequency was simulated. In this case, a 100 kHz in-band of LP filter was selected. Fig. 12 shows the time domain responses for LP filter when a 100 kHz sinusoidal input current with 100 A peak was applied to the filter. It was observed that 100 A peak RADIOENGINEERING, VOL. 23, NO. 2, JUNE 2014 693 0.01 0.1 1.0 10 100 1000 -60 -40 -20 0 20 40 Ia=50 A Ia=700 A Ia=500 A Ia=250 A Ia=1000 A Frequency, MHz Fig. 7. Simulated response of current gain k. Fig. 8. Simulated LP, BP, HP and BS of the proposed filter. Fig. 9. Simulated AP response of the proposed filter. 0.0001 0.001 0.01 0.1 1.0 10 100 -80 -60 -40 -20 0 20 Gain, dB Frequency, MHz Simulated (Q=1) Io=15 A Io=60 A Io=40 A Io=25 A Theoretical (Q=1) Io=15 A Io=25 A Io=40 A Io=60 A Fig. 10. Simulated frequency responses of the BP filter when Io is varied. 0.01 0.1 1.0 10 100 -80 -60 -40 -20 0 20 Frequency, MHz Gain, dB Simulated Ib1=250 A Ib1=1000 A Ib1=700 A Ib1=500 A Theoretical Ib1=250 A Ib1=500 A Ib1=700 A Ib1=1000 A Q 5 10 14 20 Q 5 10 14 20 Fig. 11. Simulated frequency responses of the BP filter when Ib1 is varied. Fig. 12. Time-domain input and output signal waveforms to demonstrate the dynamic range of the proposed filter. 694 M. KUMNGERN, F. KHATEB, P. PHASUKKIT, S. TUNGJITKUSOLMUN, S. JUNNAPIYA, ECCCII-BASED CURRENT-MODE... THD, Fig. 13. Dependence of the output harmonic distortion of LP filter on input current amplitude. Fig. 14. Simulated power consumption when the quality factor is varied using the bias current. input signal current level was possible without significant distortion. The result of the ILP total harmonic distortion (THD) analysis was approximately 1.2 %. The harmonic distortion increases rapidly if the input signal was increased beyond 200 A peak. The dependence of the THD of LP filter on input current amplitude was shown in Fig. 13. The THD was about 1.96 % when the input signal was increased to 190 µA (peak). The power consumption of the proposed filter with different natural frequencies fo (0.1 MHz, 1 MHz and 10 MHz) when the quality factor was varied using the bias current, was investigated. Fig. 14 shows the power consumption of the circuit with the bias current Io of 3.1 µA (fo = 0.1 MHz), 32 µA (fo = 1 MHz) and 340 µA (fo = 100 MHz) when the quality factor was varied from 1 to 20 using the bias current Ib2 from 50 (Q = 1) to 1000 µA (Q = 20) while constant Ib1 = 50 µA. The THD for different values of Q was also investigated. The simulated result was reported that for Q of 1, 3, 5 and 10, the THDs were 1.28, 2.4, 2.4, and 3.2 % for the input signals of 50, 30, 20 and 10 µA peak, respectively. 5. Conclusions A new four inputs and five outputs electronically tunable current-mode universal biquad filter employing only one CCCII and one ECCCII and two grounded capacitors was proposed. The use of grounded capacitor and absence from any resistors makes the proposed filter ideal for integrated circuit implementation. In addition, a new single input and five outputs can be achieved by using an additional CCCII to obtain four plus input currents. The proposed circuit possesses the following properties: (1) employment of three translinear current conveyors; (2) ability of realizing all the five standard biquadratic filtering functions, i.e. LP, BP, HP, BS and AP filters with one single topology, (3) separate electronic control of the parameters  o and Q, (4) employment of all grounded capacitors, (5) no need to impose critical-matching condition for realizing all the filter responses; (6) provision of low-input impedance, high-output impedance and low active and passive sensitivities and especially (7) offer of electronically tunable high Q-value filter. Simulation results are also given to demonstrate the effectiveness of our schemes. The simulation results obtained were found to be in good agreement with the theory. Acknowledgements This work was supported by King Mongkut’s Institute of Technology Ladkrabang (KMITL) (KMITL Research Fund to S. Tungjitkusolmun, M. Kumngern, P. 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